145301-8282-IJET-IJENS © February 2014 IJENS _{I J E N S}

1_{ Electronics Research Institute, El-Tahreer St. Dokki, Giza, Egypt }
2_{ Electronic and Communication Department, Faculty of Engineering, }

Cairo University, Giza, Egypt.

2_{ Electronic and Communication Department, Faculty of Engineering, }
Ain shams University, Cairo, Egypt

Corresponding author: [email protected].

**Abstract****— High-selectivity microstrip wideband bandpass filter **
**with two transmission zeros using DGS and PIN diode concepts is **
**proposed. The bandwidth and locations of the transmission zeros **
**can be adjusted by changing the characteristic impedance of open **
**stub and coupling coefficients of the dual mode resonators. A **
**prototype of planar wideband bandpass filter with 3-dB fractional **
**bandwidth 63% (1.84 - 3.43 GHz) is designed and fabricated. The **
**proposed filter topology employs RF PIN diodes as the switching **
**device. RF PIN diodes are used to reject band from UWB by **
**changing the location of diode. This filter can be adapted between a **
**narrowband state with a 14% bandwidth and an ultra wideband **
**state with a 63% bandwidth and compactness of up to 78% to be **
**obtained compared to other filters. This filter achieves a passband **
**reconfiguration ratio greater than 4:1. The measured and simulated **
**results indicate good performances high selectivity and second **
**harmonic suppression. **

**Index Term**** —Dual mode resonators, Even mode, Odd mode, RF **
**PIN diode, Defected Ground Structure (DGS), and Notched band. **

I. INTRODUCTION

Many existing microwave systems and applications have multifunction capabilities, meaning that there is a demand for reconfigurable filters [1]. Additionally, new applications are emerging for multi-functional RF systems with even more demanding and diverse requirements for reconfigurable filters e.g. to allow system reconfiguration from radar to communications. Consequently, reconfigurable filters are essential for future RF systems across commercial, defense and civil sectors. In all these areas reconfigurable filter technologies hold the key to controlling the spectrum of RF signals and eliminating interference and preserve the dynamic range under any signal receiving conditions. Reconfigurable filters can be realized in a variety of ways, but no matter what method of tuning used they must conserve their transmission and reflection co-efficient over the specified tuning range. Many traditional methods such as filter banks are being phased out due to reconfigurable filters having advantages such as flexibility and smaller footprints. Many tunable filters have been investigated in the past which control the center frequency and band width [1-4]. In this letter, a novel miniaturized dual-mode resonator with reconfigurable frequency bands and UWB filter with DGS is proposed. The characteristics of the proposed dual mode resonators are investigated numerically and validated experimentally.

This filter has the advantage of compact size, low insertion loss, sharp rejection and high out-of-band attenuation. The designed and fabricated filter is done on Rogers substrate (RO3010) with thickness of 0.635 mm, relative dielectric constant of 10.2, loss tangent= 0.0023 and the metallization thickness is 0.017 mm.

II. STRUCTURE OF THE DUAL MODE RESONATOR

The structure of the proposed dual-mode resonator was initially analyzed in [5-7] , which consists of a microstrip resonator with internal coupled lines and quarter-wavelength open-circuited stub, as displayed in Fig.1(a), where

*Y*

_{1},

###

_{1},

3
*Y* and

3

denote the characteristic admittances and electric lengths of the

microstrip line and open stub, respectively.

*odd*

*Y* ,

*Y*

*and*

_{even}_{2}indicate the odd- and even-mode characteristic admittances and electrical length of the coupled lines. Since the resonator is symmetrical in structure, the resonant condition can be analyzed by the classical method of even-and odd-mode excitation as listed in Table1. For odd-mode excitation, there is a voltage null along the symmetrical plane A-A'. The circuit structure is shown in Fig.1(a). Taking

###

_{1}=

###

_{2}=

###

_{3}=

###

for convenience, we can derive the input impedance (admittance) [5]. The coupling structure makes it possible for the filter to generate one finite transmission zero at the lower or upper stopband. Fig.1(b), shows the resonance frequencies of the degenerate modes. The horizontal axis is Y3, when varied from 0.02 S to 0.065 S, the resonance frequency of the even mode decreases almost linearly from 2.5 to 1.4GHz, while the resonance frequency of the odd mode is almost constant (around 1.83 GHz).(a)

(b)

### Hesham. A. Mohamed

1### , IEEE Member, Heba B. El-Shaarawy

2### , Esmat. A.F. Abdallah

1### , and Hadia El-Hennawy

3 145301-8282-IJET-IJENS © February 2014 IJENS _{I J E N S}

Fig. 1. (a) Structure of the proposed dual-mode stub-loaded resonator, (b) Resonance frequencies of the degenerate modes and transmission-zero

frequency against the different values of susceptance in Siemens Y3 (S)

(a) (b)

Fig. 2. Simulated current distributions for the two modes of the proposed resonator, (a) odd mode, (b) even mode.

TABLE1. RESONANT CONDITIONS OF THE EVEN-AND ODD-MODE EXCITATION[5].

Odd

mode _{tan(} _{)(} _{)}

))
(tan(
1
2
1
1
*Y*
*Y*
*Y*
*Y*
*jY*
*Y*
*o*
*o*
*ino* _{}
1
1
2
tan
*odd*
*eff*
*odd*
*L*
*K*
*C*
*f*
1
1
1
2
)
(
tan
Even
mode
]
[
tan
tan
]
[
tan
2
3
3
1
2
1
3
3
1
1
2
3
1
*even*
*even*
*even*
*even*
*even*
*even*
*ine*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*jY*
*Y*
3
3
1
2
1
2
2
tan
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*
*Y*

*K* *even* *even* *even*

*eff*
*even*
*L*
*K*
*C*
*f*
1
2
1
2
)
(
tan

Where C is the speed of light in free space, L1 is the length of the microstrip line, and

*eff*

denotes the effective dielectric constant of the substrate. As a result, the fundamental even-mode resonant frequency of the proposed resonator can be flexibly controlled while the fundamental odd-mode resonant frequency is unaffected. Slow-wave structure can be implemented with simple modification by etching a rectangular slot as shown in Fig. 2 incorporated in the open stub, thereby this aperture tend to decrease the quality factor of filter, thus increasing the bandwidth[7-8]. Coupling coefficient of resonator plays a key role in determining the bandwidth of a filter [9].

A full-wave EM eigenmode simulator CST 2011 ready-made software package was used to characterize the current patterns for the resonator. Fig. 2(a) and (b) show the current patterns of the resulting two fundamental eigenmodes. At the fundamental odd-mode resonant frequency, there is no current flowing on the open-circuited stub, and the open-circuited stub does not perturb the fundamental resonant current distribution in Fig.2 (a). On the other hand, at the fundamental even-mode resonant frequency, there is current flowing on the open-circuited stub, which changes the current distribution path, thus changing the resonant frequency in Fig. 2(b). The normalized element values of the Chebyshev low-pass filter prototype with 0.01 dB ripple can be obtained from [9] as, g0 = 1, g1 = 0.4488, g2 = 0.4077, g3 = 1.1007 that improves the passband and out-of-band performances [7]. The initial dimensions can be chosen using the above design procedure and then the CST 2011 is used to optimize the dimensions for 1.83GHz GSM applications.

III. DUAL-MODE COUPLED LINE RESONATOR

It can be seen that the line width was chosen to give the characteristic impedance of 50Ω and coupled line length and open stub are quarter wavelengths at mid frequency 1.83 GHz of GSM applications of the symmetrical structure as shown in Fig.3 (a).

In addition, the impacts of some parameters on frequency characteristics of the filter are analyzed in detail as shown in Fig.3 (b). A result, the fundamental even-mode resonant

frequency of the proposed resonator can be flexibly controlled while the fundamental odd-mode resonant frequency is unaffected. Besides, it can be seen from Fig. 3(b) that there is a transmission zero inherently through the stopband. The transmission-zero frequency of the resonator can also be tuned by varying the value of Y3. As Y3 increases, the transmission zero will shift from the upper to lower stopband. This unique property allows an easy design of asymmetric responses with improved selectivity below or above the passband, by just

varying the width

of the

open-circuited stub (W3)

as listed in Table2.

(a)

(b)

Fig. 3. (a) The geometric description of the BPF (all dimensions in mm). (b) Frequency responses of the filter ageist width of the open-circuited stub W3. We can predict and obtain the transmission zero from the analysis of the transmission coefficient S21 [7]. The simulated and measured the return loss of the filter is 0.17 dB and insertion loss is 30dB at 1.83 GHz, while the insertion loss is 26 dB, and return loss is 0.9 at 1.88 GHz. The response exhibits a transmission zero located at 2.25 GHz and a bandwidth of 270 MHz (14.5%) as shown in Fig4(a). The fabricated filter is depicted in Fig. 4(b).

145301-8282-IJET-IJENS © February 2014 IJENS

(a) (b) Fig. 4. (a) CST EM-simulated and measured S-parameters of BPF. (b) Photo of

fabrication.

TABLE II

INFLUENCE OF VARYING OPEN STUB WIDTH (W3) IN FIG.4(A)

(W3) (mm) 2 2.5 5.5 7 8 11
*A.* f*o(GHz) *

1.91 1.81 1.77 1.7 1.69 1.61

*B.* *FBW (%)* 21 14 10 8 11 19

Trans. Zero(GHz) 2.47 2.2 2.1 1.46 1.3 1.2

IV. UWBBPFDESIGN USING DGS AND RECONFIGURABLE FREQUENCY BAND

The above bandpass filters, has one transmission zero, low sharpness factors and its bandwidth is limited to 14.5%, which is only suitable for narrow-band applications. To overcome this limitation, a defected ground structure (DGS) has been proposed and implemented in the fabrication. The advantages of using the DGS under the microstrip line is that it is possible to increase the characteristic impedance by additional effective inductance generated by the DGS [4], and suppression of the second harmonic[11]. The broadened width of the DGS microstip line can be understood as the increased equivalent capacitance, which plays a great role in raising the phase constant and slow-wave effects.

(a) (b)

Fig. 5. (b) Configuration of the microstrip UWB-BPF. (all dimensions in mm). (b) Configuration of the proposed DGS.

UWB transmission systems are characterized by using an instantaneous bandwidth greater than 500 MHz or a fractional bandwidth of more than 20%. In Fig.5, we have presented a novel DGS. To illustrate this, the even-mode frequency is adjusted by d, when d is increasing, the even-mode frequency shifts higher with a ﬁxed odd-mode frequency. As discussed

above, two transmission zeros resulting from the dual-mode
resonator and the DGS coupling line can be found, as shown in
Fig.5, DGS microstrip line is used to diminish the higher order
harmonics of 2*f*o, at 3.8GHz.

Under the considerations of compact size, sharp rejection band and high performance filter, a novel BPF was designed by the DGS geometries. Simulated and measured results are shown in Fig. 6(a). It shows that the measured results matches closely the simulation ones, The results show that the proposed BPF has high a good performance with respect to the 3 dB bandwidth 1.84 - 3.43 GHz (FBW 63%), insertion loss of -0.25 dB and sharpness (the transition band between stop-band and pass-band is only 0.048GHz from 1.7982GHz to 1.8466GHz, the transition band between pass-band and stop-band is only 0.0666 GHz from 3.4329 GHz to 3.4995 GHz), and the sharpness is 351dB/GHz in low frequency edge and 255 dB/GHz in high frequency edge. UWB BPF has three transmission poles in the passband and two transmission zeros in the lower and upper stopbands to enhance selectivity. The experimental results show excellent agreement with the theoretical simulation results. Measurements were done using Agilent 7819ES Network Analyzer with range 50MHz to 13.5GHz.

(a)

(b)

Fig. 6. UWB BPF using DGS, (a) Simulated and measured S-parameters, (b) S-parameters at D1 on state.

145301-8282-IJET-IJENS © February 2014 IJENS _{I J E N S}

we have considered the configuration shown in Fig.5(b), in which three RF PIN diodes are able to guarantee a proper bandwidth of the operating frequency. The positions of the RF switches have been chosen by inspecting the path of the currents on the ground surface to individuate the most suitable placement of the diodes to guarantee the current flow.

This electronically reconfigurable filter is done mainly by RF PIN diodes (D2, D3, and D4) ON and OFF states in Fig.5 (b), DGS has two distinct characteristics of slow wave effect in passband and distinct stopband properties.

As it is known, any defect etched in the ground plane of the microstrip line disturbs its current distribution. This will increase the effective capacitance and inductance. Hence, by selectively switching PIN diode in DGS ON and OFF or altering the overall impedance of the filter changes the passband width. The simulated results with reconfigurable bandwidth are 18% at center frequency 1.77GHz for ON state which achieves a passband reconfiguration ratio greater than 4:1 as in Fig. 7(a).

(a)

(b)

Fig. 7. S-parameters results for the narrow BPF at (D2, D3, and D4) ON state, (b) Fabricated reconfigurable filter with DGS and RF PIN diode. In order to allow this structure to generate a narrow notched band inside the ultra-wide band as shown in Fig. 5(a), we found that the loading RF PIN diode (D1) in internal coupled lines in distance (K) to changes the length of the quarter wavelength to generate tuned notched bands to avoid interference of UMTS at 2.1GHz, and the group delay as demonstrated in Fig.7 (a). Fig. 7(b) illustrates the fabricated filter with bias lines to apply bias voltages to the circuit using HPND 4005 PIN diodes. The optimal positions of the RF PIN diodes are obtained on cut and trail basis. The simulated performance results of the filter in Fig.5 are summarized in Table III.

TABLE III

OBSERVED RESULTS FROM THE RECONFIGURABLE FILTER IN FIG (5) PIN diodes

state

Filter Type

S11 (dB)

S21 (dB)

FBW (3dB)

Passband (MHz)

Trans. zero All diodes off UWB

BPF

-25 -0.25 63% 1586 two

D1 ON

D2&D3&D3

OFF

UWB with notched band

-0.9 -27 2% 30 two

D2&D3&D3 On

and D1 OFF

Narrow band BPF

-35 -0.15 18% 370 one

V. RESULTSANDDISCUSSION

The overall dimension of the filter is 26 mm (length) x15 mm (breadth) x0.635 (height) mm, and the efficient electric size of the proposed filter is 0.41

###

*x0.24*

_{g}###

*with size reduction 78% relative to Table.4 and the fractional bandwidth of the UWB bandpass filter is about 63 %. The performances of the proposed filter along with the parameters of other BPFs in published literatures are compared in Table 4.*

_{g}TABLE IV

COMPARISON AMONG VARIOUS UWB BPFS. Ref.

Par.

[3] [5] [8] [10] [12] This work

Permittivity 10.2 3.38 3.5 2.65 2.2 10.2 Thickness(mm) 1.27 0.813 1.52 0.5 0.5 0.635 Loss tangent 0.0023 0.0027 0.0018 0.0025 0.002 0.0023

*f*o (GHz) 2 1.84 2 2 3 1.83

FBW 5.4% 23% 22.% 10% 43% 63%

Trans. zero One Two One One Six Two Size (mm2) 36x16 52x31 94x29 115x19 51x30 26x15 Size reduction 20% 75% 78% 82.8% 90% --- Publication year 2006 2011 2012 2009 2013

VI. CONCLUSION

145301-8282-IJET-IJENS © February 2014 IJENS

properties of compact and miniature sizes, low passband insertion losses and high frequency selectivity.

REFERENCES

[1] J.-S. Hong, “Reconfigurable planar filters”, IEEE Microwave Magzine, Vol.10, No.6, pp.73-83, Oct. 2009.

[2] Z. Birto, I. Limas and S. Colpo, “Precise frequency and bandwidth control of switchable microstrip bandpass filters using diode and microelectro-mechanical system technologies,” IET Microw. Antennas Propag, Vol. 9, No. 4, pp. 713-717, 2012.

[3] H. Zhang and K. Chen "A microstrip bandpass filter with an electronically reconfigurable transmission zero," European Microwave Conference, pp.653-656, 9-12 Sept. 2006.

[4] M. Hesham, H. El-Sharawy, E. Abdualla and H. El-Hennawy" Design of reconfigurable miniaturized UWB-BPF with tuned notched band" Progress In Electromagnetics Research B, Vol. 24, pp.97-109, 2013.

[5] C. Hua, C. Miao "Microstrip bandpass filters using dual-mode resonators with internal coupled lines," Progress In Electromagnetics Research, Vol. 21, pp.99-111, 2011.

[6] M. Matsuo and M. Makimoto, "Dual-mode stepped- impedance ring resonator for bandpass filter applications," IEEE Trans. Microw. Theory Tech., Vol. 49, No. 7, pp.1235-1240, 2001.

[7] R. Mongia, I. Bahl, and P. Bhartia, "RF and Microwave Coupled line Circuits, " Artech House, Norwood, MA, 2007.

[8] W. Apisak, and S. Kunthphong “Stepped-impedance coupled resonators for implementation of parallel coupled microstrip filters with spurious band suppression,” IEEE Trans. Microw. Theory Tech., Vol. 60, No.6, pp. 1540– 1548, June. 2012.

[9] D. M. Pozar, Microwave Engineering, 4th Ed. JohnWiley Inc., 2012. [10] C. Wen and W. Guang "Effective design of novel compact fractal-shaped

microstrip coupled-line bandpass filters for suppression of the second harmonic", IEEE, Microwave and Wireless Comp. Lett., Vol.19, No.2, pp. 74- 76, July 2009.

[11] HPND 4005, Avago Technologies, United States.

[12]S. J. Xue, W. J. Feng, and W. Q. Che,"Microstrip wideband bandpass filter with six transmission zeros using transversal signal interaction concepts" Progress In Electromagnetics Research C, Vol. 34, pp.1-12, 2013.

BIBLIOGRAPHIES

**Hesham A. Mohamed **received a BSc. degree in
Electronics and communication engineering from
the University of Menofia in 2003 and received
his M.Sc. degree from Ain Shams University in
2009. His M.Sc. is Miniaturization Techniques for
Microstrip Filters. He is currently working toward
the Ph.D. degree in the area of design and
implementation of compact and reconfigurable
planar filters at Ain Shams University. His
research interests include the design and analysis
of microstrip filters, microstrip antennas, and its
application in wireless communication. He now
holds an assistant research at the Electronics
Research Institute (ERI) institute, Giza, Egypt.

**Heba B. El-Shaarawy** was born in Cairo,
Egypt, in 1981. She has graduated from
Electronics and Communication Dept., Faculty
of Engineering, Cairo University, Egypt in 2003
with honor, and worked as a teacher assistant in
the same department. She has obtained her
master degree in 2005 from Electronics and
Communications Dept., Cairo University, in the
miniaturization of microstrip filters, and Ph.D.
degrees in 2009 from the University of
Toulouse, Toulouse, France. Her fields of
interest are microstrip components and
antennas, electromagnetic bandgap structures,
and defected ground structures.

**Esmat A. Abdallah** graduated from the Faculty of
Engineering and received the M.Sc. and Ph.D.
degrees from Cairo University, Giza, Egypt, in
1968, 1972, and 1975, respectively. She was
nominated as Assistant Professor, Associate
Professor and Professor in 1975, 1980 and 1985,
respectively. In 1989, she was appointed President
of the Electronics Research Institute ERI, Cairo,
Egypt, a position she held for about ten years. She
became the Head of the Microstrip Department,
ERI, from 1999 to 2006. Currently, is the
Microstrip Department, Electronics Research
Institute, Cairo, Egypt. She has focused her
research on microwave circuit designs, planar antenna systems and nonreciprocal
ferrite devices, and recently on EBG structures, UWB components and antenna
and RFID systems. She acts as a single author and as a coauthor on more than
160 research papers in highly cited international journals and in proceedings of
international conferences in her ﬁeld.