### Soft-Switching Current-Fed Push–Pull Converter for 250-W AC Module Applications

*Young-Ho Kim, Soo-Cheol Shin, Jung-Hyo Lee, Member, IEEE, Yong-Chae Jung,*
*and Chung-Yuen Won, Senior Member, IEEE*

**Abstract—In this paper, a soft-switching single-inductor push–**

**pull converter is proposed. A push–pull converter is suitable for**
**low-voltage photovoltaic ac module systems, because the step-up**
**ratio of the high-frequency transformer is high, and the number**
**of primary-side switches is relatively small. However, the conven-**
**tional push–pull converter does not have high efficiency because of**
**high-switching losses due to hard switching and transformer losses**
**(copper and iron losses) as a result of the high turn ratio of the**
**transformer. In the proposed converter, primary-side switches are**
**turned ON at the zero-voltage switching condition and turned OFF**
**at the zero-current switching condition through parallel resonance**
**between the secondary leakage inductance of the transformer and**
**a resonant capacitor. The proposed push–pull converter decreases**
**the switching loss using soft switching of the primary switches. In**
**addition, the turn ratio of the transformer can be reduced by half**
**using a voltage-doubler of secondary side. The theoretical analysis**
**of the proposed converter is verified by simulation and experimen-**
**tal results.**

**Index Terms—Current-fed push–pull converter, photovoltaic****(PV) ac module, soft-switching.**

NOMENCLATURE

*v*_{in} Input voltage of PV module.

*v**L bst* Voltage of boost inductor.

*i**L bst* Current of boost inductor.

*v**S 1* Voltage of switch 1.

*i**S 1* Current of switch 1.

*v**S 2* Voltage of switch 2.

*i**S 2* Current of switch 2.

*v**T 1* Voltage of upper primary transformer.

*v**T 2* Voltage of lower primary transformer.

*v**T 3* Voltage of secondary transformer.

*v**L k* Voltage of leakage inductance.

*i** _{L k}* Current of leakage inductance.

*v** _{C r}* Voltage of resonant capacitor.

*i** _{C r}* Current of resonant capacitor.

Manuscript received November 7, 2012; revised February 7, 2013 and March 17, 2013; accepted April 15, 2013. Date of current version August 20, 2013.

This work was supported by the National Research Foundation (NRF) of Korea under grant funded by the Korea government (MEST) (2011-0015584). Rec- ommended for publication by Associate Editor T. Shimizu.

Y.-H. Kim, S.-C. Shin, J.-H. Lee, and C.-Y. Won are with the School of Infor- mation and Communication Engineering, Sungkyunkwan University, Suwon 440-746, Korea (e-mail: tomeito@skku.edu; funkeee@skku.edu; daumin@

naver.com; woncy@skku.edu).

Y.-C. Jung is with the Department of Electronic Engineering, Namseoul Uni- versity, Cheonan 331-707, Korea (e-mail: ychjung@nsu.ac.kr).

Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2013.2258942

*i**V D* Current of voltage-doubler.

*V**o1* Voltage of upper voltage-doubler capacitor.

*i**o1* Current of upper voltage-doubler capacitor.

*V**o2* Voltage of lower voltage-doubler capacitor.

*i**o2* Current of lower voltage-doubler capacitor.

*V**o* Voltage of output capacitor.

I. INTRODUCTION

**D**

UE to the serious problems of environmental pollution
and fossil fuel exhaustion, photovoltaic (PV) energy has
received great attention as a source of renewable energy. Many
kinds of inverter circuits and corresponding control methods
for PV generation systems have been widely studied [1]–[4]. In
the past, PV generation systems based on centralized inverters
were mainly utilized, interfacing a large number of PV modules
to the grid. The PV modules are divided into series or parallel
connections through string diodes for high power generation.
These systems have severe limitations, such as the use of high- voltage dc cables between the PV modules and the inverter, mismatch loss between PV modules, losses in the string diodes, and nonflexible design because of mass production. String-type inverters using PV modules in series can be used to address these issues. There are no losses associated with string diodes, and separate maximum power point tracking (MPPT) can be applied to each string [5]. These increase the overall efficiency compared to the centralized inverter. However, ac-module-type inverters integrated into each PV module have received attention recently [6], [7]. This type removes the mismatch between PV modules with individual MPPTs. Each output terminal of an individual ac module inverter is connected to the ac grid. Thus, only ac cable wiring is needed, which simplifies the installation.

Extension of the system is easy due to the modular structure, and plug-and-play devices make installations easy [8].

This paper is organized as follows. The characteristics of the conventional push–pull converter are reviewed in Section II.

Modeling and implementation of the proposed soft-switching current-fed push–pull converter is presented in Section III. Sim- ulation results are illustrated in Section IV, and the experimen- tal results are presented in Section V. Conclusions are given in Section VI.

II. CONVENTIONALPUSH–PULLCONVERTER

The push–pull converter is utilized in PV ac module systems because of the advantages of fewer components, simplicity, and isolation between the PV modules and the ac grid line [9]–

[14]. In the primary side of the transformer of the conventional

0885-8993 © 2013 IEEE

TABLE I

COMPARISON OF THECONVENTIONALPUSH–PULLCONVERTER AND THEPROPOSEDPUSH–PULLCONVERTER

push–pull converter, a voltage-fed source is used. But this is not suitable for ac module systems, which should have high step-up ratio due to the low input voltage and the high output voltage.

Therefore, the push–pull converter is used with a current source,
which can decrease the turn ratio of the transformer. Since the
primary voltage of the transformer is larger than the source
voltage input by a boost inductor, the copper loss and leakage of
transformer elements can be reduced. Moreover, the converter
*needs only a small decoupling capacitor of about 10–20 μF at the*
output of the dc–dc converter. The dc link voltage and output
current are controlled at the pulse width modulation (PWM)
inverter of the two-stage configuration. Thus, the voltage ripple
of the capacitor can be slightly influenced by voltage ripple with
the frequency of the ac grid. The current-fed converter is more
appropriate for renewable energy systems than the voltage-fed
converter.

In the secondary side of the transformer, the conventional push–pull converter is designed using center-tap or full-bridge types. In the center-tap type, the voltage stress across diodes is higher than the voltage stress across diodes in the full-bridge type. Therefore, the center-tap type is not suitable for a topology with a high secondary voltage of the transformer. In the full- bridge type, the voltage stress of the diodes can be reduced because four diodes are used. Thus, the full-bridge type is more appropriate for grid-connected PV ac modules.

III. PROPOSEDPUSH–PULLCONVERTER

*A. Characteristics of the Proposed Converter*

Table I shows the comparison of the conventional push–pull converter and the proposed push–pull converter.

The conventional push–pull converter also has switching losses due to hard switching. Transformer losses also occur due to high step-up ratio to connect the low voltage of the PV module to the high voltage of the ac grid line. The main circuit of the conventional push–pull converter is presented in Fig. 1.

The main circuit of the proposed push–pull converter is shown
in Fig. 2. The circuit is composed of primary switching de-
*vices (S*_{1}*, S*_{2}*), a boost inductor L*_{bst}, a voltage-doubler, and
*an LC resonant tank. The proposed converter utilizes the soft-*
*switching technique using a parallel LC resonance [15]–[17].*

The switching losses can be reduced with the soft switching

Fig. 1. Main circuit of the conventional push–pull converter.

Fig. 2. Main circuit of the proposed push–pull converter.

when the primary switches are turned ON and OFF. The diodes
*(D*1*, D*2) of the voltage-doubler become off-state at the zero-
current switching (ZCS) condition. The transformer losses are
reduced, because the boost inductor and the voltage-doubler
decrease the turn ratio of the transformer.

*The primary-side voltage (v*_{T 1}*, v** _{T 2}*) of the transformer is

*the sum of the input voltage v*

_{in}

*and L*

_{bst}

*voltage v*

*. The*

_{L bst}*voltage-doubler voltage (V*

_{o1}*, V*

*) is half of the output voltage*

_{o2}*V*

*.*

_{o}*The secondary-side voltage v** _{T 3}*of transformer is

*v**T 3**= v**L**k* *+ v**C r* (1)
*where v** _{L k}*is the voltage of leakage inductance in the secondary

*transformer and v*

*is the voltage of resonant capacitor.*

_{C r}Fig. 3. Key waveforms of the proposed converter.

*The primary-side voltage v** _{T 1}*of transformer can be expressed
as

*v**T 1* = 1

*nv**T 3* = 1

*n(v**L**k* *+ v**C r*) (2)
*where n is the transformer turn ratio.*

*The voltage of L*bstis presented by

*v*_{L}_{b s t} *− v*in = 1

*n(v*_{L}_{k}*+ v*_{C r}*) .* (3)
*For soft switching and MPPT, both the duty ratio D and*
*switching frequency f*_{sw} should be controlled at the same time.

*B. Operation of the Proposed Converter*

Fig. 3 shows key waveforms of the proposed converter. The
*interval t*0*−t*10describes the various stages of operation during
one switching period. The converter operation is repetitive in
the switching cycle. One complete switching cycle is divided
into ten modes. The operation modes of the proposed converter
with current flows are shown in Fig. 4. The current flows have
five repetitive modes operating in the opposite direction every
switching period [7]–[9].

*Mode 1 [t*_{0}*–t*_{1}*]: Current flows through S*_{1}. The flow of the
*resonant current at the resonant tank is reversed. i** _{L k}*decreases
linearly.

*The current of L*_{bst} is

*i**L*b s t*(t) =* *v*_{in}

*L*bst

*t + I**o**,* *i**L*b s t*(t*1*) = I*1 (4)

*where I**o* *is the current of L*bst *in t*0 *time and I*1 is the current
*of L*bst*in t*1time.

*The current of L**k* can be expressed as
*i**L**k**(t) =−V**O**/2*

*L**k*

*t + i**L k**(t**o**) ,* *i**L k**(t*1*) = 0.* (5)
*The voltage of C**r*is presented by

*v**C r**(t) =* *V**O*

2 *,* *v**C r**(t*1) = *V**o*

2 *.* (6)

*As a result, i**S 1* *increases operating ZCS at turn-on and i**S 2*

decreases shaping a resonant curve. The flux already formed in
*the core of the transformer by S*2 is offset by the reverse flux
*by the turned-on S*1. Magnetic energy cannot be transferred to
the secondary side of the transformer. As the leakage current
*is decreased, it outputs to D*1 and becomes zero. The diode is
turned OFF operating ZCS without reverse recovery loss.

*Mode 2 [t*1*–t*2*]: i**S 1**increases and i**S 2*decreases shaping reso-
nant curves. The current of the resonant tank becomes negative.

*This mode is finished when i** _{S 2}*reaches zero.

*The current of L*_{bst}is

*i**L*b s t*(t) =* *v*_{in}

*L*bst

*t + I*1*,* *i**L*b s t*(t*2*) = I*2 (7)
*where I*2 *is the current of L*bst*in t*2time.

*The current of L**k* can be expressed as
*i**L k**(t) =−V**o**/2*

*Z**r*

*sin ω**r**t,* *i**L k**(t*2) =*−I**r 2* (8)
*where I**r 2* *is the current of L**k* *in t*2 time.

In (8), Z*r*is given by
*Z**r* =

*L*_{k}*C**r*

*.* (9)

*In (8), ω**r*is presented by
*ω** _{r}*= 1

*√L*_{k}*C*_{r}*.* (10)

*The voltage of C**r*is as follows:

*v**C r**(t) =V**o*

2 *cos ω**r**t,* *v**C r**(t*2*) = V**r 2* (11)
*where V**r 2* *is the voltage of C**r**in t*2time.

*Mode 3 [t*2*–t*3*]: The negative i**S 2*flows through the antiparal-
*lel diode of S*2*. In this mode, S*2is turned OFF in the zero-voltage
switching (ZVS) condition. The current of the resonant tank is
decreased to the minimum and increased again. This mode is
*finished when the magnetic energy in L*bststarts to decrease and
is released as current.

*The current of L*bstis

*i**L*b s t*(t) =* *v*in

*L*_{bst}*t + I*2*,* *i**L*b s t*(t*3*) = I*3 (12)
*where I*3 *is the current of L*bst*in t*3time.

*The current of L**k* can be expressed as
*i*_{L k}*(t) =−V*_{o}*/2*

*Z**r*

sin

*ω*_{r}*t + sin*^{−1}*I*_{r 2}*Z*_{r}*V**o**/2*

*, i*_{L k}*(t*_{3}) =*−I**r 3*

(13)
*where I*_{r 3}*is the current of L*_{k}*in t*_{3} time.

Fig. 4. Operation modes of the proposed converter with current flow.

*The voltage of C**r* is presented by
*v**C r**(t) =V**o*

2 cos

*ω**r**t + cos*^{−1}*V**r 2*

*V**o**/2*

*,* *v**C r**(t*3) =*−V**r 3*

(14)
*where V**r 3**is the voltage of C**r* *in t*3time.

*Mode 4 [t*_{3}*–t*_{4}*]: As S*_{2} is turned OFF, magnetic flux in the
*core is formed. C** _{r}*is charged by the current of the transformer.

*This mode is over when C** _{r}*is charged completely. At this time,
power is transferred to the output through the diodes of the
voltage-doubler.

*The current of L*bst is

*i**L*b s t*(t) = I*3*,* *i**L*b s t*(t*4*) = I*4 (15)

*where I*4*is the current of L*bst *in t*4 time.

*The current of L**k* can be expressed as
*i*_{L k}*(t) =* *v*_{L k}

*L**k*

*t− I**r 3**,* *i*_{L k}*(t*_{4}) =*−I**r 4* (16)

*where I*_{r 4}*is the current of L*_{k}*in t*_{4}time.

*The voltage of C**r*is presented by
*v**C r**(t) =−I**r 3*

*C*_{r}*t− V**r 3**,* *v**C r**(t*4) =*−V**o*

2 *.* (17)
*K** _{L}* is given as follows:

*K**L* =*L*_{m 2}*L**k*

(18)
*where L**M* is the magnetizing inductance in the secondary-side
transformer.

*The voltage of L**k* is presented by
*v**L k* =*−* *L**k*

*L*_{k}*+ L*_{m 2}*v**C r* =*−* 1

*1 + K*_{L}*v**C r**.* (19)
*Mode 5 [t*_{4}*–t*_{5}*]: After C*_{r}*is discharged completely, i** _{L k}*flows

*to D*

_{2}

*and transfers power to the voltage-doubler. i*

*decreases linearly and the induced secondary current of the transformer*

_{L bst}*decreases slowly. This mode is ended when S*

_{2}is turned ON.

*The equations of i*_{L bst}*and i** _{L k}* are as follows:

*The current of L*bst is

*i*_{L}_{b s t}*(t) =* *v*_{in}*− (v**C r**+ v*_{L k}*) /n*

*L*_{bst} *t + I*_{3}*,* *i*_{L}_{b s t}*(t*_{5}*) = I*_{5}
(20)
*where I*_{5}*is the current of L*_{bst} *in t*_{5} time.

*The current of L** _{k}* can be expressed as

*i*

*L k*

*(t) =−v*

_{L k}*L**k*

*t− I**r 4**,* *i**L k**(t*5) =*−I**r 5* (21)
*where I**r 5**is the current of L**k* *in t*5time.

*The voltage of C**r*is presented by
*v**C r**(t) =−V**o*

2 *,* *v**C r**(t*5) =*−V**o*

2 *.* (22)

In modes 6–10, the converter operates oppositely.

*Due to the voltage–second balance condition of L*bst, the
*voltage transfer ratio G**v* is proportional to the turn ratio. Fig. 5
*shows the voltage–second balance condition of L*bst*. G**v*is cal-
culated as follows:

*G** _{v}* =

*v*

_{o}*v*in

= *2n*

1*− D* = *2n*

1*− (f*sw*/f*res)*.* (23)
The voltage transfer ratio is proportional to the transformer
*turn ratio and the switching period T*sw. To maintain the out-
*put voltage V**o* *as the input voltage v*in changes, the switching
*frequency f*sw *should be varied. Fig. 6 presents G**v* according
*to the transformer turn ratio n. The resonant frequency f*res is
assumed to be 400 kHz. When the transformer turn ratio is 5.5,
*G**v**is sufficient to boost the maximum power point voltage v*m pp

*of 30.9 V to the output voltage V**o*of 400 V.

*Fig. 7 shows the voltage transfer ratio G**v* according to reso-
*nant frequency f*_{res}*. At this step, the turn ratio n is assumed to*
*be 5.5. G*_{v}*becomes higher with lower f*_{res}, but the losses with
*large leakage inductance are increased. With high f*_{res}sufficient
*G** _{v}* cannot be secured. So, 400 kHz is the suitable value for

*f*

_{res}[12], [13].

Fig. 5. *Voltage–second balance condition of L*b s t.

Fig. 6. *Voltage transfer ratio according to the turn ratio n.*

Fig. 7. *Voltage transfer ratio according to the resonant frequency f*re s.

Fig. 8. Control method for the duty ratio and switching frequency.

*C. Switching Implementation*

Fig. 8 shows a control method for the duty ratio and switch- ing frequency according to the variation of PV power. For soft switching over the entire range of solar radiation, Perturbation and Oscillation MPPT and a PI controller are utilized.

The resonant frequency is designed according to the voltage
*gain G**v**and transformer turn ratio n in Figs. 7 and 8. In general,*
*the switching frequency f*sw is decided in 10–20% of resonant
*frequency f*_{res}by the rule of thumb when the resonant push–pull
converter using variable frequency is designed.

The reference magnitude is determined from the switching
*period T*_{sw} *and resonance period T*_{res}*. The on-time T*_{on} of the
*switch is the sum of half of T*sw *and a quarter of T*res. After half
*of T*sw *has elapsed, since S*1 *is turned ON, S*2 is turned ON.

*After a quarter of T*res *has passed, since S*2 *is turned ON, S*1

should be turned OFF for soft switching [10], [11].

*The on-time T*on of the primary switches is given by the
following equation:

*T*on= *T*sw

2 +*T*res

4 *.* (24)

TABLE II

PARAMETERS OF THESIMULATION ANDEXPERIMENT

TABLE III

PV SIMULATORPARAMETERSACCORDING TO THEIRRADIATION

*By dividing (24) into T*sw*, D can be calculated as follows:*

*D =*1
2 + *T*_{res}

*4T*sw

*.* (25)

Fig. 9 presents a control block diagram of the dc–dc converter
and dc–ac inverter implemented using a microcontroller (DSP
TMS320F28035). In ac modules based on dc-link, the dc–dc
converter should perform MPPT [18]–[22]. The dc-link voltage
and the grid current are controlled by the dc–ac inverter. To out-
*put the maximum power, the sensed v*_{pv}*and i*_{pv}are controlled
by MPPT. The output of the MPPT is the PV reference switch-
*ing period T*_{sw}* ^{∗}* . The output is gained by a PI controller, and the

*output of the PI controller is calculated to T*on by the reference calculator. The calculated reference is compared with the out- put s of carrier calculator through dc–dc converter PWM signal

*generator. Thus, the main switches S*1

*and S*2are switched using

*these P WM*

*S 1*

*, P WM*

*S 2*.

Also, the secondary-side dc–ac inverter is controlled by mi-
*crocontroller (DSP TMS320F28035). The phase angle θ of grid*
*voltage v*gridis calculated by phase-locked loop using the sensed
*v*_{grid}. The PWM reference of dc–ac inverter is calculated by
*PI controller using the sin θ and I*_{grid}* ^{∗}* . The PWM reference is
compared with the carrier. The DC–AC inverter signal gener-
ator compares the PWM reference with carrier. The switches

*S*

_{3}

*−S*6 of the dc–ac inverter are controlled by PWM signals gained through dc–ac inverter signal generator.

Fig. 9. Control block diagram of the dc–dc converter and dc–ac inverter using a microcontroller.

Fig. 10. (a) Carriers and a reference for PWM. (b) Waveforms of primary
*switch S*1*. (c) Waveforms of primary switch S*2.

Fig. 11. *(a) Boost inductor current i**L b s t**. (b) Resonant capacitor voltage v**C r*.

Fig. 12. (a) Waveforms of tracking the MPP. (b) PWM according to MPPT. (c) Start flag of MPPT.

IV. SIMULATIONRESULTS

The proposed soft-switching push–pull converter is verified by a simulation and an experiment. The parameters of the sim- ulation and experiment are shown in Table II. The simulation is performed using PSIM 9.0. Table III shows the SC current, MPP current, OC voltage, and MPP voltage values in response to the output from an LPC250SM solar cell module. The experiment was conducted with an Agilent E4350B Solar array simulator, the details of which are shown in Table III.

Fig. 10(a) shows the carriers and the reference for PWM.

*Fig. 10(b) shows the waveforms of G**S 1**, i**S 1**, and v**S 1**of S*1. The
*waveforms of G**S 2**, i**S 2**, and v**S 2* *of S*2 are shown in Fig. 10(c).

The turn-on and turn-off under ZCS and ZVS conditions are,
respectively, confirmed from the waveforms of primary switches
*S*_{1}*and S*_{2}. The switches are turned OFF when the current flows
through the antiparallel diode.

*i*_{L bst}*has a 200-mA ripple current, and v** _{C r}* swings from

*−V**o**/2 to V** _{o}*/2, as shown in Fig. 11. Fig. 12(a) shows the tracking
operation for the maximum power point. The carriers and the

*reference varied by MPPT are shown in Fig. 12(b). Higher f*sw

is needed to track the maximum power point. The start flag of MPPT is presented in Fig. 12(c).

V. EXPERIMENTALRESULTS

In order to control the proposed push–pull converter, a TMS320F28035 microcontroller is used (Texas Instruments).

The input voltage of the PV simulator is about 20–40 V. For the

Fig. 13. *Current and voltage waveforms of switch S*1 according to ZVS and
ZCS.

initial charge of the dc link capacitor, a soft starting method is used.

*L**k*is adjusted according to winding length of the transformer.

By regulating the space between the primary- and secondary- side wires, the secondary leakage inductance is adjusted, and the degree of coupling is changed. Because of the large primary current, a rectangular compacted Litz wire is used.

Fig. 13 shows experimental waveforms of the main switch
*S*_{1}*. When the gate signal G*_{S 1}*is applied to S*_{1}*, the S*_{1}is turned
ON with ZCS and turned OFF with ZVS.

*The resonant capacitor voltage v** _{C r}*and boost inductor current

*i*

_{L bst}*are shown in Fig. 14. The magnitude of the v*

*and*

_{C r}*the ripple of i*

*are confirmed. Fig. 15 shows the efficiency*

_{L bst}Fig. 14. Waveforms of resonant capacitor voltage and boost inductor current.

Fig. 15. Efficiency of the proposed converter.

of the proposed converter. The maximum efficiency is 96.6%

according to the load conditions.

VI. CONCLUSION

In this paper, the soft-switching single-inductor push–pull converter for PV ac module applications is proposed. Soft switching was confirmed at each part, and MPPT is per- formed for extracting the maximum power from the PV module.

Switches of the primary side operate in the ZVS condition at
turn-off and in the ZCS condition at turn-on. The proposed con-
*verter maintains a V**o*of 400 V to provide ac 220 Vrm sfor dc–ac
inverters. The maximum efficiency is 96.6%. These results were
confirmed by simulation and verified by a 250-W experimental
setup.

REFERENCES

[1] S. B. Kjaer, J. K. Pedersen, and F. Blaabjerg, “A review of single-phase
*grid-connected inverters for photovoltaic modules,” IEEE Trans. Ind.*

*Appl., vol. 41, no. 5, pp. 1292–1306, Sep. 2005.*

[2] S. B. Kjaer, J. K. Pedersen, and F. Blaabjerg, “Power inverter topologies
*for photovoltaic modules: A review,” in Proc. IEEE. Ind. Appl. Conf.,*
vol. 2, Oct. 2002, pp. 782–788.

[3] Y. Xue, L. Chang, S. B. Kjaer, J. Bordonau, and T. Shimizu, “Topolo-
gies of single-phase inverters for small distributed power generators: An
*overview,” IEEE Trans. Power Electron., vol. 19, no. 5, pp. 1305–1314,*
Sep. 2004.

[4] R. Gonzalez, J. Lopez, P. Sanchis, and L. Marroyo, “Transformerless in-
*verter for single-phase photovoltaic systems,” IEEE. Trans. Power Elec-*
*tron., vol. 22, no. 2, pp. 693–697, Mar. 2007.*

[5] T. Shimizu, K. Wada, and N. Nakamura, “Flyback-type single-phase utility
interactive inverter with power pulsation decoupling on the DC input for an
*AC photovoltaic module system,” IEEE Trans. Power Electron.,, vol. 21,*
no. 5, pp. 1264–1272, Sep. 2006.

[6] D. Meneses, F. Blaabjerg, O. Garc´ıa, and J. A. Cobos, “Review and com-
parison of step-up transformerless topologies for photovoltaic AC-Module
*application,” IEEE Trans. Power Electron., vol. 28, no. 6, pp. 2649–2663,*
Jun. 2013.

[7] A. C. Nanakos, E. C. Tatakis, and N. P. Papanikolaou, “A weighted-
efficiency-oriented design methodology of flyback inverter for AC photo-
*voltaic modules,” IEEE Trans. Power Electron., vol. 27, no. 7, pp. 3221–*

3233, Jul. 2012.

[8] A. C. Kyritsis, E. C. Tatakis, and N. P. Papanikolaou, “Optimum design of
the current-source flyback inverter for decentralized grid-connected pho-
*tovoltaic systems,” IEEE Trans. Energy Convers., vol. 23, no. 1, pp. 281–*

293, Mar. 2008.

[9] W. C. P. de Aragao Filho and I. Barbi, “A comparison between two current-
fed push-pull DC-DC converter-analysis design and experimentation,” in
*Proc. IEEE. Telecommun. Energy Conf., Oct. 1996, pp. 313–320.*

[10] D. A. Ruiz-Caballero and I. Barbi, “A new flyback-current-fed push-pull
*DC-DC converter,” IEEE Trans. Power Electron., vol. 14, no. 5, pp. 1056–*

1064, Nov. 1999.

[11] E.-H. Kim and B.-H. Kwon, “High step-up resonant push-pull converter
*with high efficiency,” IET Trans. Power Electron., vol. 2, no. 5, pp. 79–89,*
Jan. 2009.

[12] L. V. Hartmann, M. B. R. Correa, and A. M. N. Lima, “A simple and
effective control strategy for improved operation of a current-fed push-
*pull converter,” in Proc. IEEE. Energy Convers. Congr. Expo., Sep. 2010,*
pp. 1098–1103.

[13] T. C. Lim, B. W. Williams, S. J. Finney, H. B. Zhang, and C. Croser,

*“Energy recovery snubber circuit for a dc-dc push-pull converter,” IET*
*Trans. Power Electron., vol. 5, no. 6, pp. 863–872, Jul. 2012.*

[14] S. M. Mukhtar, A. R. M. Saad, and N. H. Hanafi, “A high efficiency
microcontroller-based step-up push-pull DC-DC converter for PV in-
*verter,” in Proc. IEEE. Power Energy Eng. Conf., Dec. 2010, pp. 141–145.*

[15] M. J. Ryan, W. E. Brumsickle, D. M. Divan, and R. D. Lorenz, “A new
*ZVS LCL-resonant push-pull DC-DC converter topology,” IEEE Trans.*

*Ind. Appl., vol. 34, no. 5, pp. 1164–1174, Oct. 1998.*

[16] E. Adib and H. Farzanehfard, “Zero-voltage transition current-fed full-
*bridge PWM converter,” IEEE. Trans. Power Electron., vol. 24, no. 4,*
pp. 1040–1047, Apr. 2009.

[17] C.-L. Chu and C.-H. Li, “Analysis and design of a current-fed zero-
voltage-switching and zero-current-switching CL-resonant push-pull
*dc-dc converter,” IET Trans. Power Electron., vol. 2, no. 4, pp. 456–465,*
Jul. 2009.

[18] K. Ishaque, Z. Salam, M. Amjad, and S. Mekhilef, “An improved particle
swarm optimization (PSO)–based MPPT for PV with reduced steady-state
*oscillation,” IEEE. Trans. Power Electron., vol. 27, no. 8, pp. 3627–3638,*
Aug. 2012.

[19] N. Kasa, T. Iida, and L. Chen, “Flyback inverter controlled by sensorless
*current MPPT for photovoltaic power system,” IEEE Trans. Ind. Electron.,*
vol. 52, no. 4, pp. 1145–1152, Aug. 2005.

[20] M. A. G. de Brito, L. Galotto, L. P. Sampaio, G. de Azavedo e Melo, and
C. A. Canesin, “Evaluation of the main MPPT techniques for photovoltaic
*applications,” IEEE. Trans. Ind. Electron., vol. 60, no. 3, pp. 1156–1167,*
Mar. 2013.

[21] B. Subudhi and R. Pradhan, “A comparative study on maximum power
*point tracking techniques for photovoltaic power systems,” IEEE Trans.*

*Sustainable Energy, vol. 4, no. 1, pp. 89–98, Jan. 2013.*

[22] A. K. Abdelsalam, A. M. Massoud, S. Ahmed, and P. N. Enjeti,

“High-performance adaptive perturb and observe MPPT technique for
*photovoltaic-based microgrids,” IEEE. Trans. Power Electron., vol. 26,*
no. 4, pp. 1010–1021, Apr. 2011.

**Young-Ho Kim was born in Korea in 1981. He re-**
ceived the M.S. degree in the Graduate School of
Photovoltaic System Engineering, Sungkyunkwan
University, Suwon, Korea, in 2010, where he is cur-
rently working toward the Ph.D. degree.

His research interests include converters, invert- ers, and their control for photovoltaic applications.

**Soo-Cheol Shin was born in Korea in 1977. He**
received the B.S. degree in electrical engineering
from Suwon University, Hwaseong, Korea, in 2004,
and the M.S. degree in energy system engineer-
ing from Sungkyunkwan University, Suwon, Korea,
in 2006, where he is currently working toward the
Ph.D. degree in electronic electrical and computer
engineering.

He is also a Senior Researcher of Power & In- dustrial System R&D Center, Hyosung Corporation, Seoul, Korea. His research interests include robust control, wind power system, dc distribution system, and high power system including multilevel and HVDC.

**Jung-Hyo Lee (S’08–M’13) was born in Korea in**
1982. He received the B.S. degree in electrical en-
gineering from Konkuk University, Seoul, Korea, in
2006, and the M.S. and the Ph.D. degrees in electrical
engineering from Sungkyunkwan University, Suwon,
Korea, in 2008 and 2013, respectively.

His research interests include converters and in- verters for motor drive applications.

**Yong-Chae Jung was born in Korea in 1965. He**
received the B.S. degree from Hanyang University,
Seoul, Korea, in 1989, and the M.S. and Ph.D. de-
grees in electrical engineering from the Korea Ad-
vanced Institute of Science and Technology, Daejeon,
Korea, in 1991 and 1995, respectively.

He is currently an Associate Professor in the Department of Electronic Engineering, Namseoul University, Cheonan, Korea. His research interests in- clude the design and control of power converters, soft- switching power converters, resonant power circuits, photovoltaic systems, power factor correction, switched-mode power supply, induction heating circuits, and electromagnetic-interference suppression.

Dr. Jung is a member of the Korea Institute of Power Electronics and the Korea Institute of Electrical Engineers.

**Chung-Yuen Won (SM’05) was born in Korea in**
1955. He received the B.S. degree in electrical en-
gineering from Sungkyunkwan University, Suwon,
Korea, in 1978, and the M.S. and Ph.D. degrees in
electrical engineering from Seoul National Univer-
sity, Seoul, Korea, in 1980 and 1987, respectively.

From 1990 to 1991, he was with the Department of Electrical Engineering, University of Tennessee, Knoxville, as a Visiting Professor. Since 1988, he has been with a member of the faculty of Sungkyunkwan University, where he is a Professor in the College of Information and Communication Engineering. He is also the Director of Samsung Energy Power Research Center. He was the President of the Korean Institute of Power Electronics in 2010. Since 2011, he has been the Director of the Korean Federation of Science and Technology Societies. His current research interests include the power electronic of electric machines, electric/hybrid ve- hicle drives, and power converters for renewable energy systems.