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Soft-Switching Current-Fed Push–Pull Converter for 250-W AC Module Applications

Young-Ho Kim, Soo-Cheol Shin, Jung-Hyo Lee, Member, IEEE, Yong-Chae Jung, and Chung-Yuen Won, Senior Member, IEEE

Abstract—In this paper, a soft-switching single-inductor push–

pull converter is proposed. A push–pull converter is suitable for low-voltage photovoltaic ac module systems, because the step-up ratio of the high-frequency transformer is high, and the number of primary-side switches is relatively small. However, the conven- tional push–pull converter does not have high efficiency because of high-switching losses due to hard switching and transformer losses (copper and iron losses) as a result of the high turn ratio of the transformer. In the proposed converter, primary-side switches are turned ON at the zero-voltage switching condition and turned OFF at the zero-current switching condition through parallel resonance between the secondary leakage inductance of the transformer and a resonant capacitor. The proposed push–pull converter decreases the switching loss using soft switching of the primary switches. In addition, the turn ratio of the transformer can be reduced by half using a voltage-doubler of secondary side. The theoretical analysis of the proposed converter is verified by simulation and experimen- tal results.

Index Terms—Current-fed push–pull converter, photovoltaic (PV) ac module, soft-switching.


vin Input voltage of PV module.

vL bst Voltage of boost inductor.

iL bst Current of boost inductor.

vS 1 Voltage of switch 1.

iS 1 Current of switch 1.

vS 2 Voltage of switch 2.

iS 2 Current of switch 2.

vT 1 Voltage of upper primary transformer.

vT 2 Voltage of lower primary transformer.

vT 3 Voltage of secondary transformer.

vL k Voltage of leakage inductance.

iL k Current of leakage inductance.

vC r Voltage of resonant capacitor.

iC r Current of resonant capacitor.

Manuscript received November 7, 2012; revised February 7, 2013 and March 17, 2013; accepted April 15, 2013. Date of current version August 20, 2013.

This work was supported by the National Research Foundation (NRF) of Korea under grant funded by the Korea government (MEST) (2011-0015584). Rec- ommended for publication by Associate Editor T. Shimizu.

Y.-H. Kim, S.-C. Shin, J.-H. Lee, and C.-Y. Won are with the School of Infor- mation and Communication Engineering, Sungkyunkwan University, Suwon 440-746, Korea (e-mail: tomeito@skku.edu; funkeee@skku.edu; daumin@

naver.com; woncy@skku.edu).

Y.-C. Jung is with the Department of Electronic Engineering, Namseoul Uni- versity, Cheonan 331-707, Korea (e-mail: ychjung@nsu.ac.kr).

Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2013.2258942

iV D Current of voltage-doubler.

Vo1 Voltage of upper voltage-doubler capacitor.

io1 Current of upper voltage-doubler capacitor.

Vo2 Voltage of lower voltage-doubler capacitor.

io2 Current of lower voltage-doubler capacitor.

Vo Voltage of output capacitor.



UE to the serious problems of environmental pollution and fossil fuel exhaustion, photovoltaic (PV) energy has received great attention as a source of renewable energy. Many kinds of inverter circuits and corresponding control methods for PV generation systems have been widely studied [1]–[4]. In the past, PV generation systems based on centralized inverters were mainly utilized, interfacing a large number of PV modules to the grid. The PV modules are divided into series or parallel connections through string diodes for high power generation.

These systems have severe limitations, such as the use of high- voltage dc cables between the PV modules and the inverter, mismatch loss between PV modules, losses in the string diodes, and nonflexible design because of mass production. String-type inverters using PV modules in series can be used to address these issues. There are no losses associated with string diodes, and separate maximum power point tracking (MPPT) can be applied to each string [5]. These increase the overall efficiency compared to the centralized inverter. However, ac-module-type inverters integrated into each PV module have received attention recently [6], [7]. This type removes the mismatch between PV modules with individual MPPTs. Each output terminal of an individual ac module inverter is connected to the ac grid. Thus, only ac cable wiring is needed, which simplifies the installation.

Extension of the system is easy due to the modular structure, and plug-and-play devices make installations easy [8].

This paper is organized as follows. The characteristics of the conventional push–pull converter are reviewed in Section II.

Modeling and implementation of the proposed soft-switching current-fed push–pull converter is presented in Section III. Sim- ulation results are illustrated in Section IV, and the experimen- tal results are presented in Section V. Conclusions are given in Section VI.


The push–pull converter is utilized in PV ac module systems because of the advantages of fewer components, simplicity, and isolation between the PV modules and the ac grid line [9]–

[14]. In the primary side of the transformer of the conventional

0885-8993 © 2013 IEEE




push–pull converter, a voltage-fed source is used. But this is not suitable for ac module systems, which should have high step-up ratio due to the low input voltage and the high output voltage.

Therefore, the push–pull converter is used with a current source, which can decrease the turn ratio of the transformer. Since the primary voltage of the transformer is larger than the source voltage input by a boost inductor, the copper loss and leakage of transformer elements can be reduced. Moreover, the converter needs only a small decoupling capacitor of about 10–20 μF at the output of the dc–dc converter. The dc link voltage and output current are controlled at the pulse width modulation (PWM) inverter of the two-stage configuration. Thus, the voltage ripple of the capacitor can be slightly influenced by voltage ripple with the frequency of the ac grid. The current-fed converter is more appropriate for renewable energy systems than the voltage-fed converter.

In the secondary side of the transformer, the conventional push–pull converter is designed using center-tap or full-bridge types. In the center-tap type, the voltage stress across diodes is higher than the voltage stress across diodes in the full-bridge type. Therefore, the center-tap type is not suitable for a topology with a high secondary voltage of the transformer. In the full- bridge type, the voltage stress of the diodes can be reduced because four diodes are used. Thus, the full-bridge type is more appropriate for grid-connected PV ac modules.


A. Characteristics of the Proposed Converter

Table I shows the comparison of the conventional push–pull converter and the proposed push–pull converter.

The conventional push–pull converter also has switching losses due to hard switching. Transformer losses also occur due to high step-up ratio to connect the low voltage of the PV module to the high voltage of the ac grid line. The main circuit of the conventional push–pull converter is presented in Fig. 1.

The main circuit of the proposed push–pull converter is shown in Fig. 2. The circuit is composed of primary switching de- vices (S1, S2), a boost inductor Lbst, a voltage-doubler, and an LC resonant tank. The proposed converter utilizes the soft- switching technique using a parallel LC resonance [15]–[17].

The switching losses can be reduced with the soft switching

Fig. 1. Main circuit of the conventional push–pull converter.

Fig. 2. Main circuit of the proposed push–pull converter.

when the primary switches are turned ON and OFF. The diodes (D1, D2) of the voltage-doubler become off-state at the zero- current switching (ZCS) condition. The transformer losses are reduced, because the boost inductor and the voltage-doubler decrease the turn ratio of the transformer.

The primary-side voltage (vT 1, vT 2) of the transformer is the sum of the input voltage vin and Lbst voltage vL bst. The voltage-doubler voltage (Vo1, Vo2) is half of the output voltage Vo.

The secondary-side voltage vT 3of transformer is

vT 3= vLk + vC r (1) where vL kis the voltage of leakage inductance in the secondary transformer and vC ris the voltage of resonant capacitor.


Fig. 3. Key waveforms of the proposed converter.

The primary-side voltage vT 1of transformer can be expressed as

vT 1 = 1

nvT 3 = 1

n(vLk + vC r) (2) where n is the transformer turn ratio.

The voltage of Lbstis presented by

vLb s t − vin = 1

n(vLk + vC r) . (3) For soft switching and MPPT, both the duty ratio D and switching frequency fsw should be controlled at the same time.

B. Operation of the Proposed Converter

Fig. 3 shows key waveforms of the proposed converter. The interval t0−t10describes the various stages of operation during one switching period. The converter operation is repetitive in the switching cycle. One complete switching cycle is divided into ten modes. The operation modes of the proposed converter with current flows are shown in Fig. 4. The current flows have five repetitive modes operating in the opposite direction every switching period [7]–[9].

Mode 1 [t0–t1]: Current flows through S1. The flow of the resonant current at the resonant tank is reversed. iL kdecreases linearly.

The current of Lbst is

iLb s t(t) = vin


t + Io, iLb s t(t1) = I1 (4)

where Io is the current of Lbst in t0 time and I1 is the current of Lbstin t1time.

The current of Lk can be expressed as iLk(t) =−VO/2


t + iL k(to) , iL k(t1) = 0. (5) The voltage of Cris presented by

vC r(t) = VO

2 , vC r(t1) = Vo

2 . (6)

As a result, iS 1 increases operating ZCS at turn-on and iS 2

decreases shaping a resonant curve. The flux already formed in the core of the transformer by S2 is offset by the reverse flux by the turned-on S1. Magnetic energy cannot be transferred to the secondary side of the transformer. As the leakage current is decreased, it outputs to D1 and becomes zero. The diode is turned OFF operating ZCS without reverse recovery loss.

Mode 2 [t1–t2]: iS 1increases and iS 2decreases shaping reso- nant curves. The current of the resonant tank becomes negative.

This mode is finished when iS 2reaches zero.

The current of Lbstis

iLb s t(t) = vin


t + I1, iLb s t(t2) = I2 (7) where I2 is the current of Lbstin t2time.

The current of Lk can be expressed as iL k(t) =−Vo/2


sin ωrt, iL k(t2) =−Ir 2 (8) where Ir 2 is the current of Lk in t2 time.

In (8), Zris given by Zr =

Lk Cr

. (9)

In (8), ωris presented by ωr= 1

√LkCr. (10)

The voltage of Cris as follows:

vC r(t) =Vo

2 cos ωrt, vC r(t2) = Vr 2 (11) where Vr 2 is the voltage of Crin t2time.

Mode 3 [t2–t3]: The negative iS 2flows through the antiparal- lel diode of S2. In this mode, S2is turned OFF in the zero-voltage switching (ZVS) condition. The current of the resonant tank is decreased to the minimum and increased again. This mode is finished when the magnetic energy in Lbststarts to decrease and is released as current.

The current of Lbstis

iLb s t(t) = vin

Lbstt + I2, iLb s t(t3) = I3 (12) where I3 is the current of Lbstin t3time.

The current of Lk can be expressed as iL k(t) =−Vo/2



ωrt + sin−1 Ir 2Zr Vo/2

, iL k(t3) =−Ir 3

(13) where Ir 3 is the current of Lk in t3 time.


Fig. 4. Operation modes of the proposed converter with current flow.

The voltage of Cr is presented by vC r(t) =Vo

2 cos

ωrt + cos−1 Vr 2


, vC r(t3) =−Vr 3

(14) where Vr 3is the voltage of Cr in t3time.

Mode 4 [t3–t4]: As S2 is turned OFF, magnetic flux in the core is formed. Cris charged by the current of the transformer.

This mode is over when Cris charged completely. At this time, power is transferred to the output through the diodes of the voltage-doubler.

The current of Lbst is

iLb s t(t) = I3, iLb s t(t4) = I4 (15)

where I4is the current of Lbst in t4 time.

The current of Lk can be expressed as iL k(t) = vL k


t− Ir 3, iL k(t4) =−Ir 4 (16)

where Ir 4is the current of Lk in t4time.


The voltage of Cris presented by vC r(t) =−Ir 3

Crt− Vr 3, vC r(t4) =−Vo

2 . (17) KL is given as follows:

KL =Lm 2 Lk

(18) where LM is the magnetizing inductance in the secondary-side transformer.

The voltage of Lk is presented by vL k = Lk

Lk + Lm 2vC r = 1

1 + KLvC r. (19) Mode 5 [t4–t5]: After Cris discharged completely, iL kflows to D2and transfers power to the voltage-doubler. iL bstdecreases linearly and the induced secondary current of the transformer decreases slowly. This mode is ended when S2 is turned ON.

The equations of iL bstand iL k are as follows:

The current of Lbst is

iLb s t(t) = vin− (vC r+ vL k) /n

Lbst t + I3, iLb s t(t5) = I5 (20) where I5is the current of Lbst in t5 time.

The current of Lk can be expressed as iL k(t) =−vL k


t− Ir 4, iL k(t5) =−Ir 5 (21) where Ir 5is the current of Lk in t5time.

The voltage of Cris presented by vC r(t) =−Vo

2 , vC r(t5) =−Vo

2 . (22)

In modes 6–10, the converter operates oppositely.

Due to the voltage–second balance condition of Lbst, the voltage transfer ratio Gv is proportional to the turn ratio. Fig. 5 shows the voltage–second balance condition of Lbst. Gvis cal- culated as follows:

Gv = vo vin

= 2n

1− D = 2n

1− (fsw/fres). (23) The voltage transfer ratio is proportional to the transformer turn ratio and the switching period Tsw. To maintain the out- put voltage Vo as the input voltage vin changes, the switching frequency fsw should be varied. Fig. 6 presents Gv according to the transformer turn ratio n. The resonant frequency fres is assumed to be 400 kHz. When the transformer turn ratio is 5.5, Gvis sufficient to boost the maximum power point voltage vm pp

of 30.9 V to the output voltage Voof 400 V.

Fig. 7 shows the voltage transfer ratio Gv according to reso- nant frequency fres. At this step, the turn ratio n is assumed to be 5.5. Gv becomes higher with lower fres, but the losses with large leakage inductance are increased. With high fressufficient Gv cannot be secured. So, 400 kHz is the suitable value for fres[12], [13].

Fig. 5. Voltage–second balance condition of Lb s t.

Fig. 6. Voltage transfer ratio according to the turn ratio n.


Fig. 7. Voltage transfer ratio according to the resonant frequency fre s.

Fig. 8. Control method for the duty ratio and switching frequency.

C. Switching Implementation

Fig. 8 shows a control method for the duty ratio and switch- ing frequency according to the variation of PV power. For soft switching over the entire range of solar radiation, Perturbation and Oscillation MPPT and a PI controller are utilized.

The resonant frequency is designed according to the voltage gain Gvand transformer turn ratio n in Figs. 7 and 8. In general, the switching frequency fsw is decided in 10–20% of resonant frequency fresby the rule of thumb when the resonant push–pull converter using variable frequency is designed.

The reference magnitude is determined from the switching period Tsw and resonance period Tres. The on-time Ton of the switch is the sum of half of Tsw and a quarter of Tres. After half of Tsw has elapsed, since S1 is turned ON, S2 is turned ON.

After a quarter of Tres has passed, since S2 is turned ON, S1

should be turned OFF for soft switching [10], [11].

The on-time Ton of the primary switches is given by the following equation:

Ton= Tsw

2 +Tres

4 . (24)





By dividing (24) into Tsw, D can be calculated as follows:

D =1 2 + Tres


. (25)

Fig. 9 presents a control block diagram of the dc–dc converter and dc–ac inverter implemented using a microcontroller (DSP TMS320F28035). In ac modules based on dc-link, the dc–dc converter should perform MPPT [18]–[22]. The dc-link voltage and the grid current are controlled by the dc–ac inverter. To out- put the maximum power, the sensed vpvand ipvare controlled by MPPT. The output of the MPPT is the PV reference switch- ing period Tsw . The output is gained by a PI controller, and the output of the PI controller is calculated to Ton by the reference calculator. The calculated reference is compared with the out- put s of carrier calculator through dc–dc converter PWM signal generator. Thus, the main switches S1and S2are switched using these P WMS 1, P WMS 2.

Also, the secondary-side dc–ac inverter is controlled by mi- crocontroller (DSP TMS320F28035). The phase angle θ of grid voltage vgridis calculated by phase-locked loop using the sensed vgrid. The PWM reference of dc–ac inverter is calculated by PI controller using the sin θ and Igrid . The PWM reference is compared with the carrier. The DC–AC inverter signal gener- ator compares the PWM reference with carrier. The switches S3−S6 of the dc–ac inverter are controlled by PWM signals gained through dc–ac inverter signal generator.


Fig. 9. Control block diagram of the dc–dc converter and dc–ac inverter using a microcontroller.

Fig. 10. (a) Carriers and a reference for PWM. (b) Waveforms of primary switch S1. (c) Waveforms of primary switch S2.

Fig. 11. (a) Boost inductor current iL b s t. (b) Resonant capacitor voltage vC r.


Fig. 12. (a) Waveforms of tracking the MPP. (b) PWM according to MPPT. (c) Start flag of MPPT.


The proposed soft-switching push–pull converter is verified by a simulation and an experiment. The parameters of the sim- ulation and experiment are shown in Table II. The simulation is performed using PSIM 9.0. Table III shows the SC current, MPP current, OC voltage, and MPP voltage values in response to the output from an LPC250SM solar cell module. The experiment was conducted with an Agilent E4350B Solar array simulator, the details of which are shown in Table III.

Fig. 10(a) shows the carriers and the reference for PWM.

Fig. 10(b) shows the waveforms of GS 1, iS 1, and vS 1of S1. The waveforms of GS 2, iS 2, and vS 2 of S2 are shown in Fig. 10(c).

The turn-on and turn-off under ZCS and ZVS conditions are, respectively, confirmed from the waveforms of primary switches S1and S2. The switches are turned OFF when the current flows through the antiparallel diode.

iL bst has a 200-mA ripple current, and vC r swings from

−Vo/2 to Vo/2, as shown in Fig. 11. Fig. 12(a) shows the tracking operation for the maximum power point. The carriers and the reference varied by MPPT are shown in Fig. 12(b). Higher fsw

is needed to track the maximum power point. The start flag of MPPT is presented in Fig. 12(c).


In order to control the proposed push–pull converter, a TMS320F28035 microcontroller is used (Texas Instruments).

The input voltage of the PV simulator is about 20–40 V. For the

Fig. 13. Current and voltage waveforms of switch S1 according to ZVS and ZCS.

initial charge of the dc link capacitor, a soft starting method is used.

Lkis adjusted according to winding length of the transformer.

By regulating the space between the primary- and secondary- side wires, the secondary leakage inductance is adjusted, and the degree of coupling is changed. Because of the large primary current, a rectangular compacted Litz wire is used.

Fig. 13 shows experimental waveforms of the main switch S1. When the gate signal GS 1 is applied to S1, the S1is turned ON with ZCS and turned OFF with ZVS.

The resonant capacitor voltage vC rand boost inductor current iL bst are shown in Fig. 14. The magnitude of the vC r and the ripple of iL bst are confirmed. Fig. 15 shows the efficiency


Fig. 14. Waveforms of resonant capacitor voltage and boost inductor current.

Fig. 15. Efficiency of the proposed converter.

of the proposed converter. The maximum efficiency is 96.6%

according to the load conditions.


In this paper, the soft-switching single-inductor push–pull converter for PV ac module applications is proposed. Soft switching was confirmed at each part, and MPPT is per- formed for extracting the maximum power from the PV module.

Switches of the primary side operate in the ZVS condition at turn-off and in the ZCS condition at turn-on. The proposed con- verter maintains a Voof 400 V to provide ac 220 Vrm sfor dc–ac inverters. The maximum efficiency is 96.6%. These results were confirmed by simulation and verified by a 250-W experimental setup.


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Young-Ho Kim was born in Korea in 1981. He re- ceived the M.S. degree in the Graduate School of Photovoltaic System Engineering, Sungkyunkwan University, Suwon, Korea, in 2010, where he is cur- rently working toward the Ph.D. degree.

His research interests include converters, invert- ers, and their control for photovoltaic applications.


Soo-Cheol Shin was born in Korea in 1977. He received the B.S. degree in electrical engineering from Suwon University, Hwaseong, Korea, in 2004, and the M.S. degree in energy system engineer- ing from Sungkyunkwan University, Suwon, Korea, in 2006, where he is currently working toward the Ph.D. degree in electronic electrical and computer engineering.

He is also a Senior Researcher of Power & In- dustrial System R&D Center, Hyosung Corporation, Seoul, Korea. His research interests include robust control, wind power system, dc distribution system, and high power system including multilevel and HVDC.

Jung-Hyo Lee (S’08–M’13) was born in Korea in 1982. He received the B.S. degree in electrical en- gineering from Konkuk University, Seoul, Korea, in 2006, and the M.S. and the Ph.D. degrees in electrical engineering from Sungkyunkwan University, Suwon, Korea, in 2008 and 2013, respectively.

His research interests include converters and in- verters for motor drive applications.

Yong-Chae Jung was born in Korea in 1965. He received the B.S. degree from Hanyang University, Seoul, Korea, in 1989, and the M.S. and Ph.D. de- grees in electrical engineering from the Korea Ad- vanced Institute of Science and Technology, Daejeon, Korea, in 1991 and 1995, respectively.

He is currently an Associate Professor in the Department of Electronic Engineering, Namseoul University, Cheonan, Korea. His research interests in- clude the design and control of power converters, soft- switching power converters, resonant power circuits, photovoltaic systems, power factor correction, switched-mode power supply, induction heating circuits, and electromagnetic-interference suppression.

Dr. Jung is a member of the Korea Institute of Power Electronics and the Korea Institute of Electrical Engineers.

Chung-Yuen Won (SM’05) was born in Korea in 1955. He received the B.S. degree in electrical en- gineering from Sungkyunkwan University, Suwon, Korea, in 1978, and the M.S. and Ph.D. degrees in electrical engineering from Seoul National Univer- sity, Seoul, Korea, in 1980 and 1987, respectively.

From 1990 to 1991, he was with the Department of Electrical Engineering, University of Tennessee, Knoxville, as a Visiting Professor. Since 1988, he has been with a member of the faculty of Sungkyunkwan University, where he is a Professor in the College of Information and Communication Engineering. He is also the Director of Samsung Energy Power Research Center. He was the President of the Korean Institute of Power Electronics in 2010. Since 2011, he has been the Director of the Korean Federation of Science and Technology Societies. His current research interests include the power electronic of electric machines, electric/hybrid ve- hicle drives, and power converters for renewable energy systems.




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