The insertion of segmentation cells in the PET and the BILBO register adds significant area overhead and degrades circuit performance. In addition, the insertion of segmentation cells in the RTL may affect the system timing due to the unequal sequential depth in the BIST mode. The synchronous sequential circuit at the RTL can be viewed as combinational circuit interconnected with storage elements. The storage elements are assumed to be edge-triggered D-type flip-flops, and the sequential circuit representation in the graph model is an acyclic graph. For example, if the BIST capabilities of G1 and G2 in Fig. 11 are removed, both G1 and G2 become normal registers (parallel-in/ parallel-out). C1, C2, and C3 can be considered as one combinational block, C, called combinational equivalence. Now, the input of C is fed from the BIST BSR input cells and the output of C feeds the BIST BSR output cells. In the BIST mode, BIST BSR input cells are configured as a TPG and BIST BSR output cells are configured as an MISR and C is tested in one test session. The test pattern, applied to the combinational equivalence, is held for d clock cycles, where d is the maximum sequential depth of the circuit. The approach to pseudo-exhaustively test the combinational circuits can also be applied to this type of the sequential circuit without timing violation and less hardware overhead. In this paper, the BIST design solution to support these types of sequential circuits, based on the BIST boundary scan architecture (IEEE-1149.1) is presented.
In this chapter, we briefly describe the steps and techniques used in a modern inte- grated circuit manufacturing process. It is not our aim to present a detailed description of the fabrication technology, which easily deserves a complete course [Plummer00]. Rather we aim at presenting the general outline of the flow and the interaction between the vari- ous steps. We learn that a set of optical masks forms the central interface between the intrinsics of the manufacturing process and the design that the user wants to see trans- ferred to the silicon fabric. The masks define the patterns that, when transcribed onto the different layers of the semiconductor material, form the elements of the electronic devices and the interconnecting wires. As such, these patterns have to adhere to some constraints in terms of minimum width and separation if the resulting circuit is to be fully functional. This collection of constraints is called the design rule set, and acts as the contract between the circuit designer and the process engineer. If the designer adheres to these rules, he gets a guarantee that his circuit will be manufacturable. An overview of the common design rules, encountered in modern CMOS processes, will be given. Finally, an overview is given of the IC packaging options. The package forms the interface between the circuit implemented on the silicon die and the outside world, and as such has a major impact on the performance, reliability, longevity, and cost of the integrated circuit.
By reducing the no of transistor count the overall switching delay, power, and area consumption can be reduced . LOW POWER CLOCKED PASS TRANSISTOR FLIP- FLOP Low Power Clocked Pass Transistor flip-flop design shows much less power & Area constraints than the Existing two Flip-Flop designs. LCPTFF will be having very less clock delay when compared to all other circuits.
Throughout most of the past history of integratedcircuits, on-chip interconnect wires were considered to be second class citizens that had only to be considered in special cases or when performing high-precision analysis. With the introduction of deep-submicron semiconductor technologies, this picture is undergoing rapid changes. The parasitics effects introduced by the wires display a scaling behaviour that differs from the active devices such as transistors, and tend to gain in importance as device dimensions are reduced and circuit speed is increased. In fact, they start to dominate some of the relevant metrics of digitalintegratedcircuits such as speed, energy-consumption, and reliability. This situation is aggravated by the fact that improvements in technology make the production of ever larger die sizes economically feasible, which results in an increase in the average length of an interconnect wire and in the associated parasitic effects. A careful and in-depth analysis of the role and the behaviour of the interconnect wire in a semiconductor technology is therefore not only desirable, but even essential.
Similar to digitalcircuits, analog and mixed-signal (AMS) circuits are also susceptible to supply-chain attacks such as piracy, over- production, and Trojan insertion. However, unlike digitalcircuits, supply-chain security of AMS circuits is less explored. In this work, we propose to perform “logic locking” on digital section of the AMS circuits. The idea is to make the analog design intentionally suffer from the effects of process variations, which impede the operation of the circuit. Only on applying the correct key, the effect of process variations are mitigated, and the analog circuit performs as desired. We provide the theoretical guarantees of the security of the circuit, and along with simulation results for the band-pass filter, low-noise amplifier, and low-dropout regulator, we also show experimental results of our technique on a band-pass filter.
The graph in Figure 4 shows the average number of correct classiﬁcation test data obtained by simula- tion of the digital circuit obtained from RRS, for the 10 experiments and for each of the 6 metrics tested. It is observed from the graph that the average number of correct answers of classiﬁcation initially increases from k = 1 independent of the metric used, and after a certain value for k the chart tends to stabilize. Best result is obtained for k = 7 and using the modiﬁed VDM metric. The worst performance of the circuit was obtained us- ing the Bayesian metric (for low values of k, k < 5) and overlap (for higher values of k, k > 5).
time . There is a great deal of difficulty concerning thermal simulation circuits of this scale, nonetheless simulation is important because thermal effects constitute the majority of failure modes as well as the major contributer to circuit aging. Newer fabrication technologies such as SOI and especially 3DIC architectures greatly magnify thermal issues due to the fact that power dissipating elements are more densely packed and physically further from the heatsink. Furthermore they are separated by higher thermally resistive materials. A number of approaches and techniques have been applied to electrothermal simulation and are covered in Section 2.7. The basics of the source of heat within an IC are presented in Section 2.2. Once heat is produced within an IC, it will flow to another region. The mechanics of this heat transfer are covered in Section 2.3. In particular, the mode of surface contact conduction is explored in Section 2.3.1 by noting the partial differential heat equation. Section 2.9 summarizes the lessons learned from this chapter.
Ternary Content Addressable Memory (TCAM) design is selected to demonstrate and evaluate the benefits of 3D IC technology in this thesis. TCAMs are regularly structured as other memory circuits do. The regularity of the circuits makes it easier to duplicate on upper tiers and reduces the design complexity associated with multiple tier design in 3D IC. TCAMs have larger and more complicated cells with additional functions compared to other memories such as SRAM or DRAM. TCAM are facing challenges in high-capacity low- power designs due to interconnects such as matchlines connecting every cell in a row and other memory characteristics that are discussed in the following sections in more details.
addition, the flip-flop circuits have positive feedback in nature, which prohibits any variation from the predefined level. Otherwise, this variation forces the output to shift either to ground or power supply voltage. The restoration circuit is modified as LATCH/RESTORER circuit by adding a pass transistor properly. Both HOLD and restore operations performed at the same time by using the circuit given in Fig. 4.
by swapping two feedback lines in a differential architecture an using an even number of inverting stages. Figure 2.11.b shows the differential version of the ring oscillator introduced explained in previous literature [8,9]. The differential or ECL design shown in Figure 2.11.b offers lower susceptibility to supply noise and easy access to outputs with quadrature phase (for RF communication). The largest disadvantage of this approach is that output amplitude is not full-swing and is dependent on the operating frequency (or bias levels). As the P and N devices are biased to starve the inverter of current, the amplitude of the output waveform will decrease. In digital implementations, additional circuitry is needed to bring the outputs to full-swing, increasing the complexity and power consumption of the circuit. Additionally, as power supply voltages are reduced, the stack-up of the devices poses a problem in the ability to bias the devices into the correct operating region. Figure 2.11.c shows the next step in the evolution of the inversion stage design . By cross coupling the PMOS load devices, much like in a differential amplifier, the output voltage swing of the circuit is increased. This design scales easier with process advancement and avoids the complex buffer designs required for the "ECL-like" designs. Nearly all modern frequency synthesis applications use some form of the differential delay stage in the VCO. The superior rejection of noise, and the accessibility to the quadrature outputs makes it more suited to the most common applications  . The choice of full-swing or partial-swing must be made with the end application in mind. Partial swing designs are easier to test in high-frequency designs, as a full-swing circuit requires a larger buffer to drive the probe/bonding pad. No one design is superior to the other, and the application and test environment should help to determine the best approach.
Mona Khanjanimoaf was born in 1991 in Bandar-e Anzali. She received the electrical engineering in 2013 from Shahid Rajaee Teacher Training University (SRTTU Tehran, Iran). Her research interest is to design Analog integrated circuit, especially the design of low-voltage and low-power circuits.
Be careful when doing the labs: The exercises in this book require the reader to strip wire and to use simple logic chips. While a young person could do the exercises in this book, it is intended for an adult audience or at least adult supervision. The parts are small, pointy and sharp, and care should be used when handling them. Clipping and stripping wires can result in small pieces of plastic and metal becoming airborne. The components used in these circuits can become very hot, especially if installed backwards. While there is nothing inherently dangerous in working with the circuits, care should be used. Safety glasses are recommended, and if any chip or part of the circuit become hot, quickly remove the power by disconnecting the battery. Do not touch any hot chips or other components, and wait for chips or other components to cool before handling them.
Equations (2.22) and (2.23) hold true along different families of characteristic curves in the x-t plane; one family corresponding to the forward or incident wave and the other to the backward or reflected wave. It is not possible for these equations to be directly integrated (Branin 1967). However if these equations are used for the case of the lossless transmission line, where R = 0 (i.e. no resistive loss in the conductor) and G = 0 (i.e. no conductance between the microstrip and the ground plane), these equations can be directly integrated, yielding exact solutions of the form A v = + Z0 Ai and A v= -Z 0 A/, where the characteristic impedance of the line Z0 = -y/L/C. This leads to an efficient algorithm that yields not only the input and output responses but also the incident and reflected waves.
In this paper we implement a simple DC model for CNTFETs already proposed by us in order to carry out static analysis of basic digitalcircuits. To verify the validity of the obtained results, they are compared with those of Wong model, resulting in good agreement, but obtaining a lighter ensuring compile and shorter execution time, which are the main characteristics to have an easy implementation in circuit simulators for CAD applications.
In [4-9] we have already proposed a compact, semi-empirical model of CNTFET, in which we introduced some improvements to allow an easy implementation both in SPICE, using ABM library, and in Verilog-A. Our model has been implemented in  to carry out static analysis of digital gates, obtaining a significant improvement compared to Wong model [11-12].
Because the only way to get a HIGH voltage level out is to have all inputs LOW, thereby cutting off all of the BJTs, this circuit functions as a NOR gate. By wire AND-ing together "N" BJT inverter circuits, an N-input NOR gate can be realized. This type of logic circuit was used in the RTL (resistor-transistor logic) commercial logic family that was briefly used in the1960s. This circuit forms the "front end" of the 74LS02 NOR gate that is still in use today.
Some of the above-mentioned MCM arithmetic concepts may in fact further improve the implementation eﬀort of the fast DFT algorithms for certain length or bit width and may be the basis for further studies. The main result of this pa- per, however, is that due to recent advances in MCM algo- rithms, Rader and chirp-z have become viable options over the conventional radix-2 FFT. This contrasts with previously accepted understanding, as expressed by Burrus and Parks [28, page 37], who state: “if implemented on digital hard- ware, the chirp-z transform does not seem advantageous for calculating the normal DFT.”