In amplified use of hybrid, monolithic microwave and millimeter wave circuits the choice of transmission line is coplanar waveguide (CPW). It is the most striking alternative to conventional used microstrip line and stripline due to its uniplanar geometry. It consist of a centre strip with two ground planes located in the same plane [1-2] i.e., on the same surface of dielectric slab as shown in Figure.1. The ground plane being on the same surface lends itself to easy mounting of circuit elements and active devices; Drilling of holes or slots through the substrate is not needed [3-6]. CPW structures are commonly used in high-speed circuits and interconnect. It offers several advantages over microstrip which are summarized in Table 1. The use of CPW in the design of circuit components and transmission lines is not yet widespread. One reason for this is due to the lack of analytical data pertaining to the characteristics of CPW [7-8].
Waveguide is a metallic structure which is used for a transmission of an electromagnetic energy between two points. There are basically two types of waveguide. A metal waveguide and a dielectricwaveguide. Metal waveguides are typically one enclosed conductor in which an insulating medium is filled while a dielectricwaveguide consists of multiple dielectrics. Hollow waveguides are widely used for Monostatic radar level measurements in the industrial environment, and satellite communications. Conventional waveguide like Rectangular waveguide has number of losses due to small corner at the end while Circular waveguide with only conduction material has number of conducting losses, insertion loss which can reduce the power of the signal and signal strength. While taking about long distance communication, need of the high power signal and strengthen signal for transmission. A dielectric is a nonconductor of electric current. Dielectric constant and finite conductivity is solution for lossless dielectric guides for zero conductivity and the hollow metallic waveguide for microwave technology [2-4] so, dielectricwaveguide is better than the circular and rectangular waveguide. Different material has different permeability, permittivity and dielectric constant which can produce better results for return loss and insertion loss.
Aperture radiators based on rectangular waveguides are widely used in microwave technology, both as individual antennas and phased arrays with close packing of elements and wide-angle scanning when operating at high levels of microwave power. In this respect, there is a need to reduce the cross-section of the elements. Open-ended waveguides filled with a dielectric are commonly used as transceiver probes or antennas. The problem of waveguide matching with the free space becomes the main design problem for this type of antennas. Diﬀerent techniques for matching the antenna to its feeding line have been reported in the literature [1–3]. In particular, in  good matching is achieved by using an additional matching layer with lower permittivity. An alternative technique is the use of air gaps between dielectric filling elements in the waveguide to achieve high performance of both the individual radiator and the array itself; this technique has been used in the frequency range 1–3 GHz [2, 3]. There are also several types of probe design [4–7] for measuring the near-field distribution in the microwave and millimeter wave ranges. Near-field microwave probes have been extensively used in non-destructive testing. For example, in  a layered medium at the end of an open-ended waveguide was used for improved sensitivity. In  the original approach to the problem of antenna-probe miniaturization for the precision EM field measurements is presented. Recently, metamaterials have been used to achieve improved near-field properties [10, 11]. The used metamaterials consist of split ring resonators or similar metallic structures. Similar metamaterials have also been used to achieve improved directivity of the far field properties of the open-ended antennae [12, 13].
Our previous results  suggest increased accuracy in de- termination of can be obtained by using thick samples. The samples used in this work were PTFE (unsintered), polystyrene and nylon. All the samples were obtained from Polypenco Engi- neering Plastics Ltd., U.K. The profile of for the PTFE sample with 50 mm 50 mm cross section and measured thickness of 6.86 mm is shown in Fig. 3 together with the polynomial curve fitting line. Also, Fig. 3 shows the effect of sample thickness on if the actual thickness is between 6.7 mm and 6.9 mm, which could result in an uncertainty as high as 7% in if the measured thickness was assumed to be accurate. The profiles of are not shown as they overlapped (when using similar thicknesses used in Fig. 3), indicating a requirement for a tight tolerance in the sample thickness.
Dielectric resonator methods constitute one of the most useful techniques for the measurement of electromagnetic material properties in the microwave frequency range. Several geometric confi- gurations are used for this purpose and, in the present paper, we consider the case of a dielectric rod enclosed in a cylindrical metallic enclosure. To carry out dielectric measurements in this sys- tem it is necessary to know the highest permittivity constant value for which the resonance condi- tion still can be attained into the cavity. Using an approach based on magnetic and electric Hert- zian potentials we have derived the set of TE and TM modes for the relevant geometry described and, then we have calculated the valid dielectricpermittivity constant range of measurements for low-loss materials in a cylindrical cavity using a simple resonance frequency condition. Finally, we present a simple application of this method in order to determine the dielectricpermittivity con- stant of heavy oil with 11 API.
BFN samples were successfully elaborated using the sol gel process, and characterized using XRD, MEB and impedance spectroscopy. Results show a crystallization in the pure perovskite phase with monoclinic symmetry, and regular and good morphology. Dielectric measurements revealed a diffuse ferro-to-paraelectric phase with relaxor character; the latter was studied and confirmed by the modified Ushino’s law. Values of the temperature of the maximum of the permittivity are lower than those reported in the literature, and we have observed a resonance phenomenon in the frequency dependence of the permittivity, with high values of the latter, which allowed determination of the piezoelectric coefficients. These coefficients are temperature dependent and, in particular, the charge coefficient shows a slight decrease with temperature. Impedance curves showed the presence of depressed semicircles in Cole-Cole diagrams signature of a distribution of relaxation times; the corresponding behavior deviates from pure Debye relaxation.
The structure of the proposed designed for SIW is shown in Fig. 2. The whole microwave system has been integrated on the same substrate without any mechanical features. A 50 Ω transmission line as characteristic impedance is connected to integrated waveguide. Mode matching is done by tapered section to transform the quasi-TEM mode of the micro-strip line into the TE 10 mode. The model of the proposed designed has been optimized in a software for the
[1–17]. The microstrip and coplanar transmission lines used as sample- cells seem to adjust better to the electromagnetic characterization of ﬁlm-shaped materials at low microwave frequencies [9–14]. They do not require great dimensions of the samples to be characterized such as the measurements in the free space [1, 2]. They do not present air gap between the sample and the conductors such as the box-shaped cells [3– 6], the open-end waveguide probe  or ﬂanged rectangular waveguides , since the microstrip and coplanar transmission lines used as sample- cells are produced onto the sample to be characterized. The principal drawback for both sample-cells is the diﬃculty to measure low-loss materials because of the metallic losses. However, they allow changing its characteristic impedance in order to propagate the quasi-TEM mode and to perform accurate measurements, modifying the width of its conductor strip.
As expected from Mie theory, the dielectric sphere is equivalent to a magnetic dipole near the magnetic resonance mode and the magnetic field is mainly localized in the sphere as shown in Figure 4(a). However, when the CTO rods are placed beside the dielectric sphere, the magnetic field distribution is changed. The magnetic field in the gap enhances greatly while that in the sphere decreases as the rod approaches the dielectric sphere. When the distance between the rod and the sphere is reduced to zero, the intensity of the magnetic resonance even decreases by about 15% from about 5.7×10 4 A/m to
determined by contacts between Al nano-particles. At the boundaries of clusters in the alternating electric field an accumulation and redistribution of free charges occurs, which changes the initial internal electric field. It is known that at low frequencies, internal electric fields are distributed accordingly conductivity and high frequen- cies—respectively the dielectricpermittivity. Therefore, the decrease ε '' and tg δ with increasing content of alu- minum nano-particles can be explained by the appearance of a relatively strong internal field in semiconductors and nano-clusters.
[see triangles therein]. Evidently, the predictions of the formula (17) (solid line, labeled with number 1 in Fig. 1) coincide with the numerical results. Notice that ε x (ω) has a zero at ω = ω 1 ≈ 0.121ω P , which indicates the top or width of the low-frequency band gap (0 < ω < ω 1 ) in the photonic band structure for modes propagating along the growth direction of the metal-dielectric superlattice (Fig. 2). Interestingly, the dispersion relation k(ω) = (ω/c) p ε x /ε 0 , calculated with the effective
visually due to the use of metal electrodes in our system. The tuning voltage in our measurement system is limited to 40 Vp-p and the cell thickness is about 28µm. The resulting field strength of 1.43 V/µm may not be high enough to achieve maximum tunability. The other reason could be the possibility of affecting accuracies at the edge of measurement frequency ranges. The data at 1GHz by Weil el. al. was measured at the edge of their measurement frequency range (0.1-1 GHz), whereas it was in the middle in our case (frequency range: 0.01-6 GHz).
Recently, several methods have been used to optimize patch antennas with varying success, such as using a dielectric substrate of highpermittivity , Defected Microstrip Structure (DMS) , Defected Ground Structure (DGS) at the ground plane  or a combination of them, and various existing optimization algorithms such as particle swarm optimization (PSO)  and genetic algorithm [5–7]. The latter is one of the global optimization algorithms that have been used widely in the past by antenna designers for the optimization of the patch shape and size in order to achieve better overall performance of the antenna.
The geometric configuration of the proposed ground plane slot resonator is shown in Fig. 1. The structure consists of a 50 Ω microstrip line on the top layer and a rectangular slot is etched in the ground plane of substrate. Two metallic plates forming a parallel plate capacitor is soldered with the ground plane across the ground slot. The low loss dielectric material could be placed between two metallic plates to increase the loading capacitance. The maximum electric field in the slot is at the centre of the slot; where parallel plates are soldered.
Continuing with our previous work  in this paper, we present an effective method that minimizes the uncertainty in the measurement of the real dielectricpermittivity of a material. As we indicated previously, the method is valid only for low loss materials in view of the nature of the exact relations between permittivity, sample and cavity dimensions, measured resonant frequency and the unloaded Q- factor for the resonant structures. Resonant methods are the preferred technique in dielectricpermittivity measurements over non-resonant measurements , in view to their higher accuracy and sensitivity. Although, recent papers have discussed the problems associated to the estimation of uncertainty in the measurement of dielectricpermittivity of materials, but only a few have proposed a systematic methodology for the reduction of the uncertainty associated to the measurement of the dielectricpermittivity using resonant cavities .
It is apparent from the salinity proﬁle of Figure 6 that there are three distinct regions. For the ﬁrst two metres of the ridge, the salinity has a mid-level average value with high variability. From 2 to 3 m the average salinity is low with low variability and the remainder of the ridge has high salinity that is highly variable. At ﬁrst glance it appears that these three layers correspond to the sail, consolidated layer and rubble, as illustrated in Figure 4, however, the authors do not provide information on the consolidated layer. One additional link between the physical model and the ice properties can be observed from the brine volume plot. Brine volume, which is a calculated quantity, exhibits the same three layers plus an extra layer at the bottom of the ridge where the value rapidly increases. The ﬁnal layer likely corresponds to the skeletal layer at the ridge bottom. The skeletal layer is a lattice of weak ice a few centimeters thick that undergoes advective transfer with the sea water.
The dielectric parameters and have been meas- ured for a nematic liquid crystal BKS/B07 in the fre- quency range of 1 kHz to 10MHz for the temperature range of 70˚C to 135˚C. Figure 3 and Figure 4 represent typical frequency dependence spectra of the real and imaginary part of the dielectricpermittivity measured for nematic sample BKS/B07 and dye doped mixtures 1 and 2. The dielectricpermittivity is found to be either con- stant or to decrease as the frequency increases [17-22] for pure sample. Lower values of at higher frequency suggest that the molecules rotate about their long mo- lecular axis . The behaviour of for the dye based mixtures 1 and 2 is similar to that of pure sample but the values of dielectric constants are higher in comparison to the corresponding values of pure sample.
Through equations available in the literature8, we can see that the combination of the magnetic permeability and the permittivity of the absorbing composite satisfying the impedance matching condition is the key to producing a high-performance microwave absorber. Specifically, for a single layer of absorber backed by a PEC [perfect electric conductor] the reflection loss and the attenuation constant are dependent on complex magnetic permeability, permittivity, and frequency. If we consider diamagnetic carbonaceous materials, microwave absorbers are due to dielectric losses. In the case of a dielectric absorbent, assuming that µ=1 - j0, the maximum reflection loss and the attenuation coefficient depend on real and imaginary part of permittivity, dielectric loss angle tangents at a given frequency8. This is why to explain the effect observed we investigated the dielectric properties of the composites studied.