Like the coherent detection for the OQPSK signal, the coherent detection for the FQPSK signal is preferable to non-coherent detection mainly due to a good BER performance. However, in some mobile channels, because of the frequency and phase offsets caused by multipath fading—such as Rayleigh fading, co-channel and adjacent channel interference, or other impairments—it is difficult, and sometimes impossible, to recover or track the carrier frequency and phase of the received
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Fig. C.13 Constellation of XPSK withA¼ 1= ffiffiffi
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signal correctly, especially in the beginning of the reception. Under such situations, the receivers with such coherent detection suffer considerable performance loss.
Even in some cases where the receivers finally synchronize their local oscillators with the carrier frequency and phase of the received signal, the receivers experience high burst errors and outages due to long acquisition times.
To solve these problems, non-coherent detection schemes such as differential detection [17] and limiter-discriminator detection [18] are the preferred counter-measures in the fading environment. Because of the robustness to both frequency and phase offsets provided by non-coherent reception systems, they have better performance in co-channel interference (CCI) and multipath fading, especially for fast fading with a large Doppler spread. For example, they can be used in the Bluetooth system, DECT system, and ZigBee system, where GFSK modulation is adopted, so that these systems have faster data recovery, lower cost, and lower implementation complexity. In addition, since they do not need the overhead to aid the carrier recovery, they can provide higher spectral efficiency and thus capacity than coherent systems. Therefore, non-coherent detection schemes are very attrac-tive for systems that require low cost and low complexity.
In fact, the FQPSK-modulated signal was not available to be non-coherently detected at the receiver until 1999 [19]. Later, the discriminator detection for FQPSK and OQPSK was investigated in [20,21]. To the author’s best knowledge, it was the first time that this non-coherent detection scheme was reported for the OQPSK-type modulation signals, including FQPSK modulation. In this section, the simulation BER of FQPSK with coherent detection will be described due to its good BER performance. For differential detection and limiter-discriminator detection for FQPSK, the interested reader can refer to [20,21].
Since FQPSK modulation is the same as OQPSK modulation, except for their different baseband waveforms, the coherent detection or demodulation used for OQPSK can also be used for FQPSK. As we have shown in Chap.4, MSK and GMSK can also be treated as a quadrature phase modulation. Thus, some carrier recovery methods, such as the reverse modulation carrier recovery introduced in Chap.4, can be used for FQPSK. A block diagram based on the reverse modulation carrier recovery shown in Fig. 4.27 can be used for coherent demodulation of FQPSK.
Usually, a pilot signal that allows the PLL to clock it first before the information-bearing signal is preferred. This pilot-aided carrier recovery scheme is very attrac-tive in time-division multiple access (TDMA) system, such as the GSM system, in which data are transmitted in burst frames and fast carrier recovery and symbol timing synchronization are required. Each frame is further partitioned into assign-able user time slots. In each slot, for example, alternating zero and one data pattern can be inserted prior to the information data for the pilot aided transmission. In the reverse-modulation–based carrier recovery, it is required for the PLL to lock its frequency and phase to the carrier frequency and phase of the received pilot signal first. Then, the received data after the pilot data are coherently detected. Mean-while, the recovered data, in turn, are used to re-modulate the following received information-bearing modulated signal.
In the coherent demodulation of FQPSK shown in Fig.4.27, a simple fourth-order Butterworth lowpass filters after the mixer is used to replace a signal correlator, or a so-called matched filter, in an optimum trellis-coded receiver for FQPSK [13]. In fact, in most practical applications, simple lowpass filters rather than correlators are preferred for their simplicity, especially in analog designs.
Actually, FQPSK performance for coherent demodulation based on a Butterworth filter is competitive with that based on a signal correlator [13] due to its simple implementation and low cost.
FigureC.16shows the recovered eye diagrams of FQPSK-B at the output of the fourth-order Butterworth lowpass filter. Due to the narrow bandwidth of the receiver channel selection filter, the Butterworth lowpass filter has large group delay variation within the bandwidth, and such group delay variation causes ISI. As a result, it degrades the system performance. Therefore, it is necessary for the receiver to have an allpass filter as a group delay equalizer to compensate for such group delay variation. It is obvious from Fig. C.16b that the compensated or equalized eyes have less ISI at the decision instants. Figure C.16c shows the experimental eye diagrams after the group delay equalizer.
FigureC.17illustrates the BER curves of FQPSK/FQPSK-B with a Butterworth filter-based receiver. These results are obtained from MATLAB simulation. We observe thatEb/Norequired by the FQPSK-B (filtered FQPSK) receiver with group delay compensation at BER¼ 104 is only about 0.2 dB more than the FQPSK (unfiltered FQPSK) receiver with group delay compensation, or only about 1.2-dB degradation compared to theoretical OQPSK performance. It is obvious that the group delay equalizer at the receiver LPF can improve BER performance by about 0.5 dB. If an optimum receiver is used for FQPSK, the BER performance of FQPSK-B with trellis-coded (Viterbi) is only about 0.6 dB inferior to the theoretical OQPSK performance [13] and 0.6 dB superior to that of FQPSK-B with Butterworth filter at the cost of increasing hardwire implementation.
Fig. C.16 Received eye diagrams of FQPSK-B at receiver LPF output:
(a) simulation before the second-order allpass filter, (b) simulation after the second-order allpass filter, and (c) hardware implementation after the second-order allpass filter at the bit rate of 270.833 kbps