HIGH-ISOLATION AND WIDE-BAND 180◦ HYBRIDS
BASED ON ELECTRONICALLY TUNABLE LUMPED-ELEMENT FILTERS
I. Gurutzeaga1, *, A. Insausti2, E. Celayeta2, and B. Sedano1
1CEIT and Tecnun, University of Navarra, Manuel de Lardiz´abal 15,
San Sebasti´an 20018, Spain
2Tecnun, University of Navarra, Manuel de Lardiz´abal 13, San
Sebasti´an 20018, Spain
Abstract—A new high isolation lumped-element 180◦ hybrid, using
electronically adjustable filters with varactor diodes, are proposed. This design is very simple and based on only two configurable low order (N = 2) filters. Due to the limited tuning frequency range of varactor diodes, maximum near-octave frequency coverage of 2.5–5 GHz was planned in the high isolation hybrid. An impressive simulated typical isolation in the range of > 60 dB was achieved. One of the typical applications of developed hybrids could be the conversion of 70 MHz IF to microwave frequencies, with broadband mixers in single-conversion converters and very high LO rejection (>60 dB).
1. INTRODUCTION
Various types of 180◦ hybrids based on transmission lines have been widely used on broadband mixer applications, because of their reduced cost and simplicity [1–4]. However, their limited isolation figure of 20 or 30 dB, generally narrow bandwidth operation, and large occupied surface in relatively low frequencies are a number of limitations to be overcome in ultra-broadband applications, such as UWB. The proposed 180◦ hybrids and baluns in [5–8] take advantage of the equivalence between transmission lines and lumped elements filters.
On the other hand, some investigations have been carried out in recent years on tunable devices, based on metamaterial transmission
Received December 30 2011, Accepted 5 September 2012, Scheduled 28 September 2012
3.55 3.60 3.65 3.70 3.75 3.80 3.85 3.90 3.95
3.50 4.00
-60 -50 -40 -30 -20 -10
-70
0
fr eq. GHz m4
SSBM R F output s pectrum (d Bm ) with LO=3 .75GHz & IF =7 0MHz
m4 freq=
dBm (HB.Vload) =-83.82GHz .98
P_LO 10.0 dBm
X1
Vdc1
Vdc3 Vdc4
X2
R2 R=50 Ohm IFpor t
Hybrid90
HYB_LO
P has eB al=3 Ga inBal=1 dB Loss= 0.8 dB -90 0 IN
ISO
Vdc2 LOpor t
HYB_RF _0º RF por t
Hybrid90
HYB_IF
P has eB al=2 Ga inBal=0.5 dB Loss= 0.5 dB
-90
0 IN IS O R1 R=50 Ohm
Figure 1. Microwave SSBM with more than 60 dB of LO suppression.
lines [4, 9, 10], which indicate feasible work of this kind of configurable devices. This article focuses on this line of tunable devices, delving into the very high isolation property, necessary for the use of designed hybrids on the single balanced mixers (SBM) of the typical DSBM (Double Side Band Modulator) or SSBM (Single Side Band Modulator) structures, in order to achieve a high rejection of the LO.
As seen in Figure 1, the adjustable high isolation SBM mixers — X1 & X2 — are inserted on a typical Hartley type phase-shift SSBM [11], in order to achieve higher suppression level of carrier than classical approaches [12]. Four DC voltages (independent between them on first approximation) are the key to achieve a carrier-rejection of > 60 dB. Moreover, sideband suppression was implemented by the input and LO 90◦ hybrids, with the balance parameters on the state-of-the-art range.
Furthermore, the high-isolation hybrids designed in Section 2 have a significant bandwidth, on the range of the octave. One of the devices is an all-lumped variant, with only concentrated components, and therefore minimum occupied surface. The designs have been developed with the aid of commercial software ADS from AgilentTM, v.2009A, and their electromagnetic planar simulator, MOMENTUM, was extensively used.
2. HYBRID DESIGN AND DEVELOPMENT
short-LPF2
N=2 StopType =short
Fpass =5.4 GHz {- t}
P4_DELTA_IN P2_OUT1
HPF2
N =2 StopType =short Fpass= 4.76 GHz {- t} TL5
F = 2.57 E = 45 Z = 70. 7
TL 6 F =2.57 E =45 Z =70.7 P3_OUT2 P1_SIGMA_IN P3_OUT2 P1_SIGMA_IN LP F5 N=2 StopType =shor tFpass= f_s GHz
LPF7 N=2 StopType=shor t Fpass=f_h GHz P4_DELTA_IN P2_OUT1 LPF6 N=2 StopType=short
Fpass=f_s GHz HPF2
N=2 Stop Type =shor t Fpass= f_l GHz
3.00 3.25 3.50 3. 75 4. 00 4. 25 4.50
2.75 4.75 -30 -25 -20 -15 -10 -5 -35 0 -60 -50 -40 -30 -20 -10 -70 0 -10 -8 -6 -4 -12 -2 freq. GHz R L a l l p o r ts Isolation P ass re s pons es a ll p o rts
Fc (GHz) 2.5 3.75 5
f_h (GHz) 2.74 4.07 5.4
f_l (GHz) 2.41 3.58 4.76
Compact dBpp
BW=50 MHz <1 <0.4 <0.25
Ultra-Compact BW=50 MHz dBpp <1.7 <0.7 <1.4
Compact solation (dB) 74 67 66
Ultra-Compact Isolation (dB) 74 66 68
(a) Compact Version
Ohm GHz Ohm GHz I m7 freq = dB(S (4,1 ))=-67. 22 6 Min
3.820 GHz m8 freq = dB(S (2,1 ))=-3.02 8
3.75 0GHz
m7 m8
m7 freq = dB(S (4,1 ))=-67. 22 6 Min
3.820 GHz m8 freq = dB(S (2,1 ))=-3.02 8
3.75 0GHz
BW = 50MHz
Isolation S22 S33 S44 S11 S21 S34
S31 S 24
Note 1
Note 1 (b) Ultra-compact Version Note 2
Figure 2. Ideal block diagrams of compact & ultra-compact hybrids,
simulation plots of compact version and resuming table. (a) The length of the transmission lines on the schematic diagram was minimised to maintain the performance. The result was a 45◦ length line lower limit of operating frequency of hybrid (2.5 GHz). (b) The fixed corner frequency (f s) of the LPF-s between ports P1 & P2 and P1 & P3 of the schematic diagram was optimised on ADS to 4.5 GHz.
type stop band configuration were possible, but low order filters are preferred because of their simplicity when adjustable filters are used to tune the characteristics on frequency [9, 10]. This approach gives a minimum 3-varactor hybrid because all 3 LPF-s are converted to tuneable filters substituting the capacitors by varactors. Furthermore, this approximation results in two different varactor models on the hybrid, because of the combination of two capacitors in parallel with the basic third order filters.
Figure 2 shows how to achieve the simplest configurable hybrid, with only two equal varactors, and consequently, the way for a more compact device, maintaining the high-isolation aim: the cancelation of two signals on the 180◦ hybrid is based on the equality of amplitude and the 180◦ phase relations between the paths P1 → P2 → P4, and P1 → P3 → P4. Then the ideal infinite isolation (S41 = 0) could be
a compact version of hybrid, and b) two equal LPF-s between ports P1 & P2 and P1 & P3, with fixed cut-off frequencies, resulting in an ultra-compact all-lumped-element version. Modifying the ratio between cut-off frequencies of P3-P4 LPF (f h) and P2-P4 HPF (f l), high isolation could be achieved on a narrow-band margin (i.e., LO frequency on up-converters), maintaining a quasi-constant pass-response between ports P1 & P2, P1 & P3, P4 & P2 and P4 & P3.
Notes 1 & 2: 1) The length of the transmission lines on the schematic diagram was minimised to maintain the performance. The result was a 45◦length line lower limit of operating frequency of hybrid (2.5 GHz). 2) The fixed corner frequency (f s) of the LPF-s between ports P1 & P2 and P1 & P3 of the schematic diagram was optimised on ADS to 4.5 GHz.
70 MHz IF up-converter, with 50 MHz BW, and approximately 2.5÷5 GHz RF output, was used as an application example: cut-off frequencies are ideally synthesized and tuned with ADS system filters, to achieve a maximum figure of isolation between 1 and 4 accesses 70 MHz up from 50 MHz pass-band. Obviously, a trade-off between pass responses and isolation level exists: the crossing point of both is on the same point ideally, and then, 70 MHz down on frequency there is a dBpp maximum error, which is quantified on the contiguous table. As seen in the graphs and table, the ultra-compact all-lumped hybrid of Figure 2 has worse pass & return loss (RL) responses,
1.75 2.00 2.25 2.50 2.75 3.00 3.25
1.50 3.50 -20 -15 -10 -5 -25 0 -60 -50 -40 -30 -20 -10 -70 0 -20 -10 -30 -3 freq. GHz
4.25 4.50 4.75 5.00 5.25 5.50 5.75
4.00 6.00 -40 -30 -20 -10 -50 0 -60 -40 -20 -80 0 -9 -6 -12 -3 Pass responses Isolation
RL sll ports
freq . GHz
Pass responses
RL sll ports
Isolation
L=4. 6 mm W=0. 15 mm
Radi us =3.658 mm Angle= 13 0 W=0. 5 mm
H=0. 5 mm D=0. 4 mm
c_ 10 p c _10 n
L=4. 2 mm W=0. 9 mm
c_ 100p P4 _DELTA_IN Vdc= 2. 65 V
c_ 10 n c_ 100p c _10 p
Vdc= 3. 03 V P 2_OUT 1
R_ 47 k
c_ 10 p c_ 100p
sm v2019_ sc7 9
L=4. 2 mm W=0. 9 mm
c_ 100p
L=1. 515 mm W=0. 15 mm Radi us =0.5 mm Angle= 180 W=0. 15 mm
L=1. 515 mm W=0. 15 mm
c_ 100p
R_ 47k _0402 Radi us =3.658 mm Angle= 13 0 W=0. 5 mm L=4. 2 mm W=0. 9 mm
c_ 100p P1 _S IGMA_IN
P3 _O UT2L=4. 2 mm W=0. 9 mm
sm v2019_ sc7 9 c_ 10 p
Fc (GHz) 2.5 3.75 5
V_HPF (V) 3.03 9.32 18.01
V_LPF (V) 2.65 8.88 17.66
dBpp (BW = 50 MHz) <1.2 <0.5 <0.3
Isolation (dB) 69 75 78
m 3 freq= dB(S (4,1 ))=-69 .4 81 Min
2.570G Hz m4 freq = dB(S (2,1 ))=-4 .3 02
2.50 0GHz
m 3 m4
m 3 freq= dB(S (4,1 ))=-69 .4 81 Min
2.570G Hz m4 freq = dB(S (2,1 ))=-4 .3 02
2.50 0GHz
m 3 freq = dB(S (4,1 ))=-78. 23 1 Min
5.070G Hz m 4 freq = dB(S (2,1 ))=-3.56 2
5.00 0GHz
m 3 m 4
m 3 freq = dB(S (4,1 ))=-78. 23 1 Min
5.070G Hz m 4 freq = dB(S (2,1 ))=-3.56 2
5.00 0GHz
BW = 50MHz S41 S22 S33 S44 S11 S21 S34 S31 S24
BW = 50MHz S41 S22
S33 S44
S11
S21 S34 S31
S24
1.75 2.00 2.25 2.50 2.75 3.00 3.25 1.50 3.50 -20 -15 -10 -5 -25 0 -60 -50 -40 -30 -20 -10 -70 0 -20 -10 -30 -3
4.25 4.50 4.75 5.00 5.25 5.50 5.75
4.00 6.00 -10 -5 -15 0 -70 -60 -50 -40 -30 -20 -10 -80 0 -9 -6 -12 -3 freq. GHz Pass responses
Isolation RL sll ports
freq. GHz
Pass responses
Isolation RL sll ports
D=0. 4 mm
c _p47_0402
sm v2019_ sc7 9 P 1_S IGMA_IN
P 3_O UT2 c _100p_0402
L=4. 2 mm
W=0. 9 mm c_ 100p_0402 l_2n7
l_2n 7
L=4. 2 mm W=0. 9 mm
l_2n7 l_2n7
L=4. 2 mm W=0. 9 mm
c _100p_0402
sm v201 9_ sc7 9
c_ 10p_0402
c _10p_0402
c_ 10 n
Vd c= 2. 4 V
R_ 47 K_HP F
c_ 10 p c _100 p
P 4_DELTA_IN R_47K _LP F
c _10p c_ 100p
c _10n
Vdc= 1. 85 V L=4. 2 mm W=0. 9 mm
c_ 100p_0402
P2 _O UT1
D=0. 4 mm
Fc (GHz) 2.5 3.75 5
V_HPF (V) 2.4 8.5 17.75 V_LPF (V) 1.85 7.6 16.36 dBpp (BW = 50 MHz) <1.5 <0.55 <0.5 Isolation (dB) 69 74 89
m 3 freq= dB(S (4,1 ))=-69 .4 81 Min
2.570G Hz m4 freq = dB(S (2,1 ))=-4 .3 02
2.50 0GHz
m 3 m4
m 3 freq= dB(S (4,1 ))=-69 .4 81 Min
2.570G Hz m4 freq = dB(S (2,1 ))=-4 .3 02
2.50 0GHz
m 3 m 4
m3 freq= dB(S (4,1 ))=-73 .0 22 Min
5.070G Hz m4 freq = dB(S (2,1 ))=-4.3165.000G Hz
BW = 50MHz S41
S22
S33
S44 S11
S21 S34
S31 S24
BW = 50MHz S41 S22 S33 S44 S11 S21 S34 S31 S24
Figure 4. Schematic & quasi-realistic simulations of ultra-compact
hybrid. V_HPF V_LPF V_HPF V_LPF (a) (b)
Figure 5. (a) Compact and (b) ultra-compact 180◦ hybrids layouts
& photos.
but their advantage is, obviously, the compactness. Both cases have less dBpp on higher frequencies, which is better to optimise the up-converter behaviour.
mounting (SMD) components, and the varactors are also standard type, SMV2019 model on SC-79 case, from Skyworks Solutions Inc [13]. It is important to remark that improving the state-of-the-art solutions [9, 10], in this case, both varactors could be the same model.
Schematic diagrams of both hybrids and the results of the quasi-realistic simulations are presented in Figures 3 and 4. The data in the tables, summarize the implementations, with an acceptable error on pass losses — less than 1.5 dBpp worst case — and a very high isolation of>60 dB.
The filters with fixed corners in Figure 4 were designed using the same inductance as the adjustable LPF and HPF, in order to minimize the number of different components.
3. IMPLEMENTED HYBRIDS AND MEASUREMENT RESULTS
Two hybrids, presented in Figure 5, were mounted on a FR-4 substrate to verify the validity of high isolation in real conditions. Two independent polarization voltages were used on varactor biasing, in order to compensate the devices impairments and easily achieve the high isolation point.
The compendium of measurements of both hybrids is presented in Figures 6 and 7, with RL & Isolation, and Pass responses, presented in separated graphics. Due to a board fabrication and components tolerances, and specially the varactors impairments, the DC controls, V HPF and V LPF presented in the tables in Figures 6 and 7, are slightly different from quasi-realistic simulations in Figures 3 and 4. This causes different cross-points between configurable HPF & LPF-s of the hybrid, and appreciable differenceLPF-s (eLPF-specially at 2.5 GHz) between simulation and measurements on RL and Pass responses. However, the utility of both high-isolation hybrids to reject high level signal, such as LO on SBM mixer with a valid pass responses dBpp, is guaranteed. To demonstrate it, the great similarity between simulated and measured high isolation (S41) and simulated and measured pass
responses are remarked.
2.1 2.2 2. 3 2.4 2. 5 2.6 2.7 2.8 2. 9
2.0 3.0
-10 -5
-15 0
-60 -50 -40 -30 -20 -10
-70 0
2.425 2.450 2.475 2.500 2.525 2.550 2.575
2.400 2.600
-7 -6 -5
-8 -4
4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4
4.5 5.5
-9 -6 -3
-12 0
-60 -40 -20
-80 0
4.92 4.94 4.96 4.98 5.00 5.02 5.04 5.06 5.08
4.90 5.10
-4.0 -3.5
-4.5 -3.0
Measu rements Realistic Simulation
Fc (GHz) 2.5 3.75 5 2.5 3.75 5
V_HPF (V) 1.29 5.62 15.32 1.3 6.79 14.72
V_LPF (V) 0.85 6.76 16.9 0.82 6.13 14.14
dBpp (BW= <1.53 <0.6 <0.6 <2.4 <1.2 <0.85
Isolation (dB) 70 8 0 7 5 7 2 7 5 8 4 Simulation Measurement
freq. GHz
RL (dB)
Measured
Pass Respo
nses (dB)
Singma to
OUTs & D
elta t
o OUTs (d
B)
Singma to
OUTs & D
elta t
o OUTs (d
B)
Isolatio
n (dB)
Isolatio
n (dB)
Simulation Measurement
Simulation Measurement Simulation Measurement
freq. GHz
freq. GHz freq. GHz
50 MHz) 5 0MHz BW
S11:
S22:
S33:
S44:
S41:
S21: S24:
S31: S34:
50 MHz BW
S41:
S21: S24:
S31: S34:
50 MHz BW 50 MHz BW
S21: S24:
S31: S34:
Figure 6. Compact 180◦ hybrid RL & isolation, and pass responses
measurements & octave-range compendium table.
In the all-lumped ultra-compact hybrid, the parasites are greater, because of higher amount of lumped components, and the higher coupling between them (e.g., lumped-inductors that have an inductive coupling because of proximity). Thus, the differences between simulated pass responses and measured ones are higher than compact version, and the range of the hybrid is less than an octave: the actual frequency range extends from 2.5 to 4.33 GHz (1.7 ratio).
Comparison between simulation and measurement is presented only on the low part of frequency range (2.5 GHz), and the high isolation property is highlighted. The table in the figure remarks the utility of hybrid to reject high level signal like LO on SBM mixer with a valid pass responses dBpp.
2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9
2.0 3.0
-15 -10 -5
-20 0
-60 -40 -20
-80 0
2.425 2.450 2.475 2.500 2.5 25 2.550 2.57 5
2.400 2.600
-6.5 -5.5 -4.5
-7.5 -3.5
3.93 4.03 4.13 4.23 4.33 4.43 4.53 4.63 4.73
3.83 4.83
-9 -6
-12 -3
-60 -40 -20
-80 0
Summary Table Measurements
Fc (GHz) 2.5 3.75 4.33
V_HPF (V) 1.89 7.55 12.36
V_LPF (V) 2.12 9.34 16.9
dBpp (BW= <2.3 <2.2 <1.6
Isolation (dB) >70 >63 >74 Simulation Measurement
freq. GHz
RL (dB)
Measured
Pass Respo
nses (dB)
Singma to
OUTs & D
elta t
o OUTs (d
B)
Isolatio
n (dB)
Isolatio
n (dB)
Simulation Measurement
Simulation Measurement
freq. GHz
freq. GHz
50 MHz) S21: S24: S31: S34: S41:
50 MHz BW
50 MHz BW
S11: S33: S22: S44:
50 MHz BW S21: S24: S31: S34:
S41:
Figure 7. Ultra-compact 180◦ hybrid RL & isolation, pass responses,
& quasi-octave range compendium table.
Ne twork Analyzer for S- pa rame te rs
hyb_ TplusLumped SP_ NWA1
St op =3 .87 GHz St art=3.77 GHz
+ +
2 1
R1 R=50 Oh m
R2 R=50 Ohm
Low level isolation (+0dBm)
High level isolation (+10dBm) considering the intermodulation introduced by
varactors Low level: 0 dBm
High level: 10 dBm
P4_DELTA_ IN
S21
S12
Figure 8. Compact Hybrid: measurement set, and comparison
“high-level”isolationof the hybrid: in other words, it is a measurement related to the intermodulation generated by the varactors non-linearity, which would affect the actual isolation when a high-level signal like LO is injected to the hybrid of the SBM mixer on the converter scheme (see Figure 1).
The plot of Figure 8 is a measurement of parameter S41 of a
compact hybrid, i.e., isolation, and was extracted with Agilent Network Analyzer E5071A, using two levels of power source: 0 dBm low-level, and +10 dBm high-level. In both cases, the two varactor polarisations were varied (V HPF and V LPF in Figures 3 & 4 & 5) to search the point of maximum isolation.
As seen in the figure, the isolation is slightly degraded, but still useful: i.e., for a typical single balanced mixer application, LO level would be in the range of +10 dBm, and then a high practical rejection would be possible, despite the contribution of varactor non-linearity.
This result proves the capability of this kind of high-rejection hybrids to form a part of the microwave frequency agile converters. The difference between S21 and S12 traces also indicates that intermodulation on varactors causes a slight non-reciprocal effect on the hybrid’s behaviour.
4. CONCLUSIONS
Two versions of compact lumped-element 180◦ hybrids, with very high and configurable isolation, were presented: one is based on a mix solution, with a pair of transmission lines and a pair of LPF and HPF; the other is fully-lumped and results in an ultra-compact design. The use of one single varactor model to cover an octave performance is quite remarkable, and the high level of practical isolation achieved is very useful for up-converter applications from 70 MHz IF to the UWB microwave band. Also, this property could be useful for direct-conversion very low-noise receivers, that need a high LO blocking, as well as many other applications.
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