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2. J.Y. Sze and K.L. Wong, Bandwidth enhancement of a microstri-pline-fed printed wide-slot antenna, IEEE Trans Antennas Propag 7 (2001), 1020–1024.
3. H.G. Schantz, A brief history of UWB antennas, IEEE Aerospace Electron Syst Mag 4 (2004), 22–26.
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7. Y. Sung, UWB monopole antenna with two notched bands based on the folded stepped impedance resonator, IEEE Antennas Wire-less Propag Lett 11 (2012), 500–502.
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omni-directional band-notch ultra-wideband antenna, Electron Lett 45 (2009), 659–660.
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A 5-GHz 4/8-ELEMENT MIMO ANTENNA
SYSTEM FOR IEEE 802.11AC DEVICES
Mohammad S. Sharawi
Department of Electrical Engineering, King Fahd University for Petroleum and Minerals (KFUPM), Dhahran 31261, Saudi Arabia; Corresponding author: [email protected]
Received 22 October 2012
ABSTRACT: A highly compact 2 2 (four element) and 2 4 (eight element) multiple-input-multiple-output (MIMO) antenna systems are designed for the IEEE 802.11ac standard. The antennas operate in the 5-GHz band with a minimum effective bandwidth of 80 MHz. The elements of the MIMO antenna system are patch antennas loaded with complementary split-ring resonators. A minimum isolation of 10.5 dB and maximum gain of0.8 dBi are measured. Total size of the MIMO antenna systems is 50 100 0.8 mm3.
VC 2013 Wiley Periodicals, Inc.
Microwave Opt Technol Lett 55:1589–1594, 2013; View this article online at wileyonlinelibrary.com. DOI 10.1002/mop.27611
Key words: multiple-input-multiple-output antennas; complementary split-ring resonator; WLAN; IEEE 802.11ac
1. INTRODUCTION
Multiple-input-multiple-output (MIMO) antenna systems have emerged as an integral part of the new 4G wireless standards.
They are a key enabling component to achieve high data rates for current and future needs of wireless communication services. Therefore, there is an increasing demand for new MIMO antenna systems which are compatible with user terminal devi-ces. In such systems, it is important to have antennas that are compact and can be easily integrated within a small form factor and yet have high isolation and low correlation factors. These requirements make the design of MIMO antenna systems challenging.
The 5-GHz band has been assigned for wireless LAN’s (WLAN) in the IEEE 802.11ac standard [1]. The standard speci-fies two-element, four-element, and eight-element MIMO anten-nas. The minimum bandwidth in this standard has been specified to be 80 MHz. This new standard will provide higher data throughputs than the current IEEE 802.11n (a maximum theoret-ical data rate of 1.733 Gbps at 80-MHz channel bandwidth and 256-QAM modulation scheme). The two main features for this new standard at the antenna side are the increase in the operat-ing bandwidth and the number of elements in the MIMO antenna system at the user terminal.
In Ref. 2, a quad-band two-element MIMO antenna was pre-sented. One of its bands of operation was 5.15–5.35 GHz. It used a k/4 resonator and a combination of C-shaped slot and T-shaped slit with a printed inverted F-antenna (PIFA). The dimensions of a unit element were 11.5 13.5 4 mm3
. Two elements were mounted on a ground plane of 50 100 mm2. The mean effective gain of each antenna element was2.1 dBi at 5.2 GHz. The antenna reported in Ref. 3 was a frequency reconfigurable two-element MIMO antenna. The antenna ele-ments of the antenna were printed monopoles. PIN diodes were used in the design for the frequency configurability. The antenna had three bands of operation out of which one was in the range of 5.15–5.35-GHz band. The dimensions of the antenna were 80 40 0.8 mm3
. The mean effective gain of the antenna ele-ments in the 5-GHz band was reported to be 1.87 dBi.
Figure 1 Geometry of the 2 4 (eight-element) MIMO antenna sys-tem; (a) top side and (b) bottom side
A dual-band MIMO antenna designed for WLAN was reported in Ref. 4. The MIMO antenna had three-elements each made up of a circular dual-loop antenna. The antenna operated in the 2.4 and 5 GHz bands. The antenna elements were placed on a circular substrate whose diameter was 120 mm. A peak gain of 8.7 dBi was reported in the 5-GHz band. In Ref. 5, a four-element dual-band MIMO antenna was pre-sented. Three different designs based on the element used were
presented. Minkowski monopole, Kochi monopole, and PIFA were used as antenna elements in the three different designs. The overall area occupied by each design was 88.7 46.6 mm2. The antenna operated in the 2.4 and 5-GHz bands. The mean effective gain of the antenna element was5.62 dBi in the 5-GHz band. A wideband, two-element MIMO antenna designed for mobile phones in the 5-GHz band was reported in Ref. 6. The antenna elements were made up of PIFA. The area
Figure 2 Fabricated MIMO antenna systems; (a) four-element and (b) eight-element. [Color figure can be viewed in the online issue, which is avail-able at wileyonlinelibrary.com]
covered by each element was 5 12 mm2
. The antenna ele-ments were placed on a ground plane whose dimensions were 50 100 1.524 mm3
. The antenna operated in the 4.7–6.2 GHz frequency range.
In this article, a compact 2 2 (four-element) and 2 4 (eight-element) MIMO antenna systems are presented. The antennas are designed to operate in the 5-GHz band to comply with the IEEE 802.11ac standard. The antenna elements consist of patch antennas loaded with complementary split-ring resona-tors (CSRRs) for antenna miniaturization. This is the first work that presents an eight-element MIMO antenna within the stand-ard user terminal size of 50 100 0.8 mm3
. The performance of the proposed MIMO antenna systems (four-element and eight-element) is measured and discussed.
The rest of the article is organized as follows. The design of the proposed MIMO antenna systems is outlined in Section 2. Simulation and measurement results are discussed in Section 3. Conclusions are summarized in Section 4.
2. 2 3 2 AND 2 3 4 MIMO ANTENNA SYSTEM DESIGN
Figure 1 shows the geometry of the proposed eight-element MIMO antenna system. The four-element design is similar but only occupies half of the top layer of the PCB. The antennas were designed on an FR-4 substrate with dielectric constant of 4.4. The thickness of the substrate was 0.8 mm. The basic antenna element was a rectangular patch with an inset feed
matched to 50X. The spacing between the elements was 5 mm. The dimensions of each patch antenna were 11 8 mm2
. Such a patch has a resonant frequency of 6.52 GHz when operated as is. A CSRR was etched out from the ground (GND) plane underneath the center of each patch antenna to reduce its reso-nant frequency.
A CSRR is the negative image of a split-ring resonator (SRR). An SRR is made up of two concentric copper rings with slits in each ring. The inner ring resides inside the outer ring with some separation while the slits in each ring are in opposite direction to one another. A CSRR is made by removing copper in the shape of an SRR from the GND plane. The CSRR inter-acts with the electric field and provides effective negative per-mittivity around its resonant frequency. The resonant frequency of a CSRR is the same as that of a SRR and it is modeled as a LC tank circuit. The resonant frequency depends on the dimensions of the CSRR. The lumped element model for a SRR and CSRR are derived in Ref. 7 which also give quasianalytical equation for finding the resonant frequency of a SRR and CSRR.
Using CSRRs over a patch or underneath, it was also investi-gated for antenna miniaturization for single antenna applications [8, 9]. A patch antenna can be modeled as a resistor-inductor-capacitor circuit at its resonance. A CSRR acts as an LC circuit. The interaction of the patch antenna and the CSRR results in an equivalent model which lowers the overall resonant frequency of the CSRR-loaded patch antenna.
In our proposed design, the dimensions of the CSRR were varied and optimized via simulations to tune the antenna at 5 GHz. The patch elements resonated at 5 GHz when the radius of the outer ring of the CSRR r was 2.5 mm, the width of the ringsw were 0.25 mm, the spacing between the rings s was 0.5 mm, and the width of the slitd was 0.5 mm. The feed line was shifted 1.5 mm along the width of patch antenna from its central axis for proper mode excitation.
3. RESULTS AND DISCUSSION
The proposed antennas were designed and optimized using HFSSTM. They were then fabricated and measured. The scatter-ing parameters of the MIMO antennas were measured usscatter-ing an Agilent HP8510C network analyzer. The two–dimensional (2D) gain measurements of the MIMO antennas were carried out at an outdoor antenna range facility (Oakland University, MI).
The dimensions of the CSRR had a profound effect on the resonant frequency of the antenna element (see Fig. 1 for pa-rameter location). It was found that d had no effect on the reso-nant frequency of the antenna. Therefore, it was kept at 0.5 mm. The resonant frequency of the antenna was found to have an inverse relationship with r. As r was increased, the resonant fre-quency of antenna decreased and vice versa. An increase in w and s resulted in an increase in the resonant frequency of antenna and vice versa. A knowledge of these relationships helped in tuning the resonant frequency of the antenna at the desired value.
Figure 2 shows the fabricated MIMO antenna systems (four-element and eight-(four-element). The measured and simulated s-pa-rameters for the four-element MIMO antenna are shown in Fig-ure 3. The resonant frequency was approximately 5.08 GHz. A minimum 6dB bandwidth of 95 MHz was measured and a minimum isolation of 10.6 dB was recorded between Elements 1 and 2. Figure 4 shows the measured reflection coefficient curves for the eight-element MIMO antenna system as well as the isolation curves between Element 1 and all others. A mini-mum bandwidth of 80 MHz was measured and minimini-mum isola-tion of 10.5 dB was obtained between Elements 1 and 2.
The evaluation of a multiport antenna system is insufficient using scattering parameters only. For proper broadband charac-terization of a multiport antenna, the total active reflection coef-ficient (TARC) should be evaluated [10]. For an N-element MIMO antenna system, TARC is given by,
Figure 6 Gain patterns of antenna Elements 1 and 2 for the 4four-ele-ment MIMO antenna measured at 5.08 GHz (a) x-z plane and (b) y-z plane. [Circles is co-pol Element 1, solid is co-pol Element 2, dots is cross-pol Element 1, and dashes are cross-pol Element 2]
C¼ ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi PN i¼1jbij 2 q ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi PN i¼1jaij2 q (1)
where ai and bi are the incident and reflected signals,
respec-tively. For the proposed antenna systems, TARC was calculated from the measureds-parameters. The amplitude of all ports was kept at unity while the excitation phases were varied with respect to Port 1. For various combinations of port excitations, TARC curves obtained for the eight-element MIMO antenna system are shown in Figure 5. For all the combinations, a mini-mum effective bandwidth of 80 MHz was achieved. Similar behavior was observed for the four-element MIMO antenna.
The envelope correlation coefficient q gives a measure of how much the antennas are correlated with each other in a MIMO antenna system. A high correlation coefficient degrades the performance of a MIMO antenna system and yields poor use of antenna diversity. The correlation coefficient can be calcu-lated from the measured scattering parameters of the antenna
system [11]. The worse case correlation values obtained were between two opposite elements, that is, 1 and 2, 3, and 4, and so forth, with a maximum value of 0.28 at 50% efficiency. This is an acceptable level of correlation within this standard. The correlation coefficient for all other antenna element pairs was less than 0.05.
The 2D gain patterns of the MIMO antennas were measured. These measurements were performed at 5.08 GHz. Figure 6(a) shows the normalized gain patterns of antenna Elements 1 and 2 of the four-element MIMO antenna system in thex-y plane, and Figure 6(b) shows the normalized gain patterns of antenna Ele-ments 1 and 2 inx-z plane. Similarly, Figure 7(a) shows the nor-malized gain patterns of antenna Elements 1 and 2 of the eight-element MIMO antenna system in thex-y plane, and Figure 7(b) shows the normalized gain patterns of antenna Elements 1 and 2 in x-z plane. The maximum gain of the antenna elements in both MIMO antenna systems was 0.8 dBi. It was observed that the gain patterns of the individual patch element were simi-lar to that of a conventional patch antenna with higher back lobes which were due to the CSRR. The low maximum gain value obtained (i.e., 0.8 dBi) was because of antenna miniaturization.
4. CONCLUSION
In this article, a highly compact 2 2 and 2 4 MIMO antenna systems were presented for the new IEEE 802.11ac standard. The design was confined within a user terminal device size of 100 50 0.8 mm3. The elements of the MIMO
anten-nas were patch antenanten-nas miniaturized using CSRR loading. The two MIMO antenna systems had an effective operating band-width of at least 80 MHz (from TARC calculations) and mini-mum isolation of 10.5 dB at 5.05 GHz. The measured gain pat-terns of the antennas of the antenna showed a maximum gain of 0.8 dBi.
ACKNOWLEDGMENTS
The author would like to acknowledge the support provided by the deanship of scientific research (DSR) at King Fahd University of Petroleum and Minerals (KFUPM), Dhahran, Saudi Arabia, through project number RG1219. Also, would like to thank Mr. M. U. Khan for his help in the simulations.
REFERENCES
1. D.A. Hall, Underneath the hood of 802.11ac, Microwave J 54 (2011), 46–52.
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3. Z.-J. Jin, J.-H. Lim, and T.-Y. Yun, Frequency reconfigurable mul-tiple input mulmul-tiple-output antenna with high isolation, IEEE Microwave Antennas Propag 6 (2012), 1095–1101.
4. S.-W. Su and C.-T. Lee, Printed, low-cost, dual-polarized dual loop-antenna system for 2.4/5 GHz WLAN access points, In: Pro-ceedings of the 5th European Conference on Antennas and Propa-gation (EUCAP), 2011, pp.1253–1257.
5. M. Karaboikis, C. Soras, G. Tsachtsiris, and V. Makios, Four-ele-ment printed monopole antenna systems for diversity and MTMO terminal devices, In: Proceedings of the 17th international Confer-ence on Applied Electromagnetics and Communications, 2003, pp. 193–197.
6. S. Vergerio, M. Elayachi, J.-P. Rossi, and P. Brachat, Design of multiple antennas at 5 GHz for mobile phone and its MIMO per-formances, In: International Conference on Electromagnetics in Advanced Applications, (ICEAA 2007), 2007, pp.17–20.
Figure 7 Gain patterns of antenna Element 1 and 2 for the eight-ele-ment MIMO antenna measured at 5.08 GHz (a) x-z plane and (b) y-z plane. [Circles is co-pol Element 1, solid is co-pol Element 2, dots is cross-pol Element 1, and dashes are cross-pol Element 2]
7. J.D. Baena, J. Bonache, F. Martin, R. Marques, F. Falcone, T. Lopetegi, M.A.G. Laso, J. Garcia, I. Gil, and M. Sorolla, Equiva-lent-circuit models for split-ring resonators and complementary split-ring resonators coupled to planar transmission lines, IEEE Trans Microwave Theor Tech 53 (2005), 1451–1461.
8. X. Cheng, et. al. Characterization of microstrip patch antennas on metamaterial substrates loaded with complementary split-ring reso-nators, Microwave Opt Technol Lett 50 (2008), 2131–2135. 9. Dong, H. Toyao, and T. Itoh, Design and characterization of
minia-turized patch antenna loaded with complementary split-ring resona-tors, IEEE Trans Antennas Propag 60 (2012), 772–785.
10. M. Manteghi and Y. Rahmat-Samii, Multiport characteristics of a wideband cavity backed annular patch antenna for multipolariza-tion operamultipolariza-tions, IEEE Trans Antennas Propag 53 (2005), 466–474. 11. P. Hallbjorner, The significance of radiation efficiencies when
using S-parameters to calculate the received signal correlation from two antennas, IEEE Antennas Wirel Propag Lett 4 (2005), 97–99.
VC2013 Wiley Periodicals, Inc.
LOW-POWER PHOTONIC CONTROL OF A
MICROWAVE RING RESONATOR USING
BULK ILLUMINATION
Mohammad Ali Shirazi-Hosseinidokht and Mani Hossein-Zadeh
Center for High Technology Materials, 1313 Goddard, SE, Albuquerque, NM 87106; Corresponding author: [email protected]
Received 22 October 2012
ABSTRACT: We demonstrate the feasibility of bulk illumination technique for low-power photonic control of RF resonance. Using this technique, the transmitted RF power through a microstripline-ring filter on a junction-less silicon substrate is changed by 11 dB with less than 2 mW of interacting optical power.VC2013 Wiley Periodicals, Inc.
Microwave Opt Technol Lett 55:1594–1599, 2013; View this article online at wileyonlinelibrary.com. DOI 10.1002/mop.27606 Key words: light-controlled RF devices; RF ring resonator; RF-photonics
1. INTRODUCTION
Photonic control of RF signal propagation is an important topic that continues to be an active area of research and development in microwave photonics [1–14]. Photonic control has many advantages over conventional electrical control. High degree of electrical isolation between the control signal and the microwave circuit, immunity to parasitic electromagnetic radiation, high power handling, overall weight reduction, high-speed control, and timing precision are among the most important benefits of photonic control. In particular, electrical isolation between the control signal and the microwave structure is crucial for design and fabrication of reconfigurable antennas [15–18] where the radiation pattern and efficiency are affected by the presence of control devices and circuits in the vicinity of antenna pattern [15].
A large variety of techniques, devices, and materials have been explored for designing photonically controlled switches, phase shifters, and attenuators. In almost all these approaches, photonic carrier generation in a semiconductor controls the am-plitude and the phase of the RF signal propagating on microstrip or coplanar transmission lines. Free carrier generation in biased and unbiased junctions as well as junction-less regions have been used to control the RF field in discontinuities [8, 11], stubs
[9, 13], resonators [6], and terminations [4, 5, 9]. Except few cases where the photosensitive element is added to a transmis-sion line fabricated on a low-loss RF substrate [1, 15, 18], in most proposed structures the RF circuit is fabricated on the pho-tosensitive semiconductor substrate in order to reduce the com-plexity of the fabrication process and keep the device mono-lithic. Although compound semiconductors have also been used as the structural materials in these devices [3, 10], implementa-tion of photonically controlled RF devices on silicon substrates is more attractive for monolithic integration of microwave and mm-wave devices using well-developed fabrication processes.
Independent of the material and the device structure, in all these approaches and devices a laser wavelength between 600 and 900 nm is used to maximize optical absorption and carrier generation. As a result, the optically affected region has been confined at the surface (due to small optical penetration depth at these wavelengths). In contrast to the previous work, here we explore the potential application of bulk illumination at a longer wavelength (1064 nm) combined with high-Q RF resonance to maximize the RF-optical field overlap. Moreover, we use a side-coupled RF ring resonator to confine the RF field and enhance the interaction of RF field and photogenerated carriers. Note that although previously certain planar resonant structures were used for photonic RF control, the confinement of free carrier on the surface has limited the interaction of the resonant field and the free carriers. In these cases, the presence of free carriers has been mainly modifying the electrical properties at the bounda-ries of the resonator (effectively tailoring the conductor size).
This article describes a photonically controlled RF ring reso-nator suitable for controlling a variety of silicon-based micro-wave integrated circuits. By optically controlling the density of free carriers in the regions of the substrate with large resonant RF field, we have demonstrated up to 8 dB of transmission loss variation with only 1.9 mW of interacting optical power from a commercial fiber pigtailed laser diode operating at 1064 nm. To our knowledge, this is the largest optical sensitivity of RF trans-mission ever reported for a passive junction-less photonically controlled microwave device.
2. PHOTONICALLY CONTROLLED RF RING RESONATOR
2.1. Bulk Versus Surface Illumination
As photonic free carrier generation controls the RF propagation in most optically controlled components, strong optical absorp-tion is the main criteria for choosing the photoconductive mate-rial and the corresponding wavelength. On the other hand to reduce the fabrication cost and complexity, preferably only one type of material is used in the device structure. As a result, low-loss optical waveguides cannot be easily integrated with the de-vice to deliver light directly to the sensitive region, and almost all devices are controlled by top illumination to avoid absorption before reaching the sensitive region. Silicon is one of the most common substrates used in phonically controlled RF devices (mainly because of compatibility with IC fabrication and low fabrication cost). Figure 1(a) shows the absorption depth (d, the depth at which the light intensity drops to 36% of its value at the interface) plotted against wavelength for silicon substrate. Below 900 lm, d is less than 30 lm, and all the photogenerated carriers are effectively confined at the surface (interface between air and silicon). That is why so far the optical control has been mainly achieved by tailoring the conductive structure on top of the semiconductor substrate using laser beam.
Here, we examine an alternative approach by choosing a wavelength with an absorption depth larger than the substrate
