plementing hundreds or thousands channels is an alter- native way of very effectively lowering the DSP speed (per channel) and still maintaining 100 Gb/s (or more) system throughput. Both Direct Detection and Coherent (heterodyne) detection is considered for OFDM imple- mentations (DD-OFDM and CO-OFDM systems) and the low channel baud-rate leads to a significant influ- ence of the laser **phase** **noise**. Especially for CO-OFDM systems the influence is severe. The theory basis for dealing with the **phase** **noise** influence has been present- ed for radio OFDM systems in [1-4] and several ac- counts for optical systems can be found in [5-9]. Using nPSK or nQAM systems with DSP based disper- sion compensation leads to strong influence of laser **phase** **noise** which is further enhanced by equalization enhanced influence of the local oscillator **phase** **noise** [10-12]. OFDM systems may use wrapping of the signal in the time domain (cyclic prefix) to account for disper- sion effects in this way eliminating the need for DSP based compensation. Using an RF carrier which is adja- cent to or part of the OFDM channel grid is an effective way of eliminating the **phase** **noise** effect [5] but it has to be noted that the dispersion influenced delay of OFDM channels will make the elimination non- complete and this leads to a transmission length de- pendent (dispersion enhanced) **phase** **noise** effect [7-9]. The purpose of this paper is to investigate this in detail for both DD-OFDM systems and CO-OFDM systems for nPSK and nQAM OFDM channel constellations using an accurate (analytical) model framework which allows direct physical insight into the problem. We will use this to derive important practical OFDM design guidelines.

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In this paper, a new approach of establishing database to estimate **phase** **noise** due to chromatic dispersion is proposed. The database can be used to compensate **phase** shift without using any pilot carriers, which results in high spectral efficiency and low BER. Time variability, temperature and aging effects of channel are introduced in order to ensure the suitability of the database for stable long term performance. The proposal is confirmed by simulation using matlab.

As shown in Fig. 3, the bulk-drain and the gate-bulk capacitances of the switching transistors, and the gate-bulk capacitances of the MOS varactors play the role of coupling devices and provide the injection path for coupling signals, thus reduce the required for any extra AC coupling capacitors and DC biasing resistors [11]. Further-more, to reduce the **noise** contribution of the core VCOs to the overall **phase** **noise**, First core VCOs are produced by 90º **phase** Shift similarly Second core VCOs are Produced. proposed QVCO Output are getting on “In **Phase** and Anti **Phase** manner” i.e., I+, I- and Q+, Q-. Due to the symmetry of the existing QVCO circuit in Fig. 2 the amplitude of all the output voltages (i.e., and ) are equal. Because of the differential structure of the two core VCOs, there is a 180º **phase** difference between the potentials of the nodes and also and . The potentials at nodes and are denoted with and in phasor domain, respectively, with is the voltage amplitude and is the **phase** difference between potentials of nodes and [1] .Next describe the QVCO result and discussion of **phase** **noise** and power.

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we showed for oscillators based on resonant microcanti- levers in [5]. This Technical **phase** **noise** due to a defi- cient **phase** control of the feedback adds to the non tech- nical, but Thermodynamical **phase** **noise** we will explain for oscillators with perfect loops where current feedback to the resonator is exactly in-**phase** with its voltage v(t) for Positive Feedback (PF) or exactly at 180˚ for Nega- tive Feedback (NF). This prevents the addition of Tech- nical **phase** **noise** to the Thermodynamical one sketched in [6] that we will explain, which is linked with resona- tor’s losses represented by its R and with **noise** added by the feedback electronics, both considered by the **noise** figure F of Leeson’s pioneering work [7]. We will show the theory behind Leeson’s empirical formula and behind the Line Broadening that these oscillators show around their mean oscillation frequency f 0 [8]. The general the-

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The authors of [21] consider a very simple special case of this problem. They investigate the accuracy of a channel estimation protocol (corresponding to protocol B1 in this work) for a two-hop network with a single source- destination pair and multiple AF relays. The gain factors are to be computed from the channel estimates at the relays in a way that all signals combine coherently at the destination antenna. The authors neglect LO **phase** **noise** and implicitly assume a perfect carrier **phase** synchronization between all relays and the destination. In comparison to [21], this work compares four diﬀerent channel estimation protocols, considers multiple source-destination pairs, takes LO **phase** **noise** into account, and drops the assumption of perfect **phase** synchronization.

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(17) From (16) and (17), we can draw the conclusion that we can reduce the **phase** **noise** of the Tx-IF or the thermal **noise** ﬂoor, or reduce both of them, to make the total output **noise** low, according to actual systems requirement. For example, if the thermal **noise** ﬂoor is dominant at the output which always happens in microwave systems, the sensitivity of radar will be improved by reducing the F of the receiver. In this case, enhancing the P Tx can enlarge the P Rx to

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will be included in F. Having considered the effects due to Dissipations of energy in the electrical **noise** of the loop, we have to consider now the effects due to each Fluctuation of electrical energy preceding each Dissipa- tion because both processes endure a charge **noise** asso- ciated with Displacement Currents (DiC) in C that dis- turb its otherwise periodic Fluctuation of charge coming from the energy exchange between magnetic and electric susceptances in an L-C resonator. The factor 2 we had to use in [1] to obtain the charge **noise** power of a series of current pulses mimicking fat TAs, reflected these two charge noises. To say it bluntly, each fast DiC of weight q in C due to a TA is followed by an opposed and slower DiC of equal weight q and opposed sense linked with its DR. This is why **phase** **noise** is a nice scenario to test the validity of the Quantum-compliant model of [2], which was able to explain 1/f excess **noise** [10] and flicker **noise** [11] as consequences of thermal **noise**.

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Abstract: In this work, we have discussed a new approach in designing the **phase**-locked loop (PLL), the proposed circuit is designed with the GDI cell-based PFD, charge pump and low pass filter. For frequency matching and for larger locking state, the design used as D flipflop from the TSMC library. It is used as frequency synthesizers and divides the incoming frequency by 2.This design uses 5 stage current starved voltage control oscillator (CS-VCO). The designed PFD is free from the dead zone issue and it is suitable for the low power applications. the designed PLL works for an average frequency range of 8 GHz and its offset frequency is targeted at 1GHz.This PLL model has low **phase** **noise** of -112 dBc/Hz at 1 Mhz frequency which is quite standard and the power consumption of the circuit is 8 µW. The entire work is simulated using Cadence Virtuoso 45 nm Technology.

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In order to reveal the **noise** contribution in the PLL based synthesizer, it is necessary to have a model in which the effect of each **noise** source into output **phase** **noise** can be analysed. The transient response of the PLL is generally nonlinear, and cannot be formulated. But, in locked condition, the PLL acts as a LTI system and hence the superposition holds. The linear model, depicted in Fig. 2, is used to analyze the behavior of the ΣΔ synthesizer in locked condition [11]. It is very useful for PLL stability and **phase** **noise** contribution analysis [12]. Note that this linearized model can be used to analyze the “small signal” dynamic properties of the PLL as well as its **noise** performance; that is, only variations in the PLL frequency caused by small changes in the divider value are considered. If the divider value is large, then cycle slip may occur and thereby invalidate our modeling assumptions [11].

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We present a comparative study on three carrier **phase** estimation algorithms, including a one-tap normalized least mean square (NLMS) method, a block-average method, and a Viterbi-Viterbi method in the n-level **phase** shift keying coherent transmission systems considering the equalization enhanced **phase** **noise** (EEPN). In these carrier **phase** estimation methods, the theoretical bit-error-rate floors based on traditional leading-order Taylor expansion are compared to the practical simulation results, and the tolerable total effective linewidths (involving the transmitter, the local oscillator lasers and the EEPN) for a fixed bit-error-rate floor are evaluated with different block size, when the fiber nonlinearities are neglected. The complexity of the three carrier **phase** estimation methods is also discussed. We find that the carrier **phase** estimation methods in practical systems should be analyzed based on the simulation results rather than the traditional theoretical predictions, when large EEPN is involved. The one-tap NLMS method can always show an acceptable behavior, while the step size is complicated to optimize. The block-average method is efficient to implement, but it behaves unsatisfactorily when using a large block size. The Viterbi-Viterbi method can show a small improvement compared to the block-average method, while it requires more computational complexity.

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The proposed design VCO is analyzed using the Cadence 180nm CMOS Technology with Virtuoso software. Figure.5. shows the transient analysis of proposed VCO. In this control voltage is set at 1V and frequency of oscillation is 2GHz.Figure.6 shows the tuning characteristics of proposed VCO. It shows the tuning range varies from 2.06 GHz to 2.62 GHz with control voltage (0.5V-1.2V). Fig. 7 represents the **phase** **noise** of proposed oscillator. It is - 112.00 dBc/Hz. This is at offset frequency of 1MHz

The input–output relationship of the fiber-optical baseband channel is described implicitly by the stochastic nonlinear Schr¨odinger equation (sNLSE) [12, Ch. 2]. It is well recog- nized that this type of channel model does not lend itself to an easy solution for various communication theoretic problems [13], [14]. We therefore consider a simplified, dispersionless channel model which follows from the sNLSE by neglecting the dispersive term and captures the interaction of Kerr- nonlinearities with the signal itself and the inline amplified spontaneous emission (ASE) **noise**, giving rise to nonlinear **phase** **noise** (NLPN) [15], [16]. A discrete channel is obtained from the waveform channel on a per-sample basis (assuming ideal carrier and timing recovery) [17, Sec. III]. This model has been previously considered by several authors in the literature and different methods have been applied to derive the joint probability density function (PDF) of the received amplitude and **phase** [3], [16]–[18]. Since all these derivations neglect dispersion, the resulting PDF should serve as a useful approximation for dispersion-managed (DM) optical links, provided that the local accumulated dispersion is sufficiently low [3, p. 160], [19]. However, if the interaction between dispersion and nonlinearities becomes too strong, the channel model is likely to diverge from the one assumed here. 1 We

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mitigate the nonlinear **phase** **noise**. By combining the return to zero (RZ) coding with modulation format at modulators, the nonlinear fiber impairments have been reduced [14,15]. Another efficient scheme called **phase**-conjugated twin waves (PCTWs) has been implemented in the optical orthogonal frequency division multiplexing (O-OFDM) and superchannel systems. In this technique the transmitted signal and its **phase**-conjugated wave are propagated though orthogonal dimensions. Co-propagating the signal with its **phase**-conjugated copy along the fiber channel can cancel the nonlinear distortions [1,3,16]. However, in PCTWs, the spectral efficiency is halved because of the 50% overhead related to transmission of the **phase**- conjugated copy of the signal [2,3].

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Xu, Tianhua, Jacobsen, Gunnar, Popov, Sergei, Li, Jie, Friberg, Ari T. and Zhang, Yimo (2011) **Phase** **noise** mitigation in coherent transmission system using a pilot carrier. In: 2011 Asia Communications and Photonics Conference and Exhibition ACP, Shanghai, China, 13-16 Nov 2011. Published in: 2011 Asia Communications and Photonics Conference and Exhibition ACP.

In this paper, an improved VCO operating at 1.5 V supply voltage and 2.33 mW low power core is proposed. The adaptive body bias and double-pseudo resistances in series with the cross-coupled pairs which are biased by the tapped node of the inductor are used to reduce the power consumption and **phase** **noise**. Measured **phase** **noise** of −122.7 dBc/Hz at 1 MHz offset from 3.07 GHz is achieved. The measured tuning range is 17.3%. The rest of this paper is organized as follows. Section 2 describes the design approach of the LC-VCO which includes the topology and circuit structure used in the design. Section 3 gives the results and analysis. Comparisons with other published works are also provided in this section. Finally, a conclusion is drawn in Section 4.

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The present paper aims are twofold. Firstly, to outline an intelligent sensor software architecture in accordance with existing standards and which provides enhanced flexibility to meet the requirements of adverse applications. Secondly, in order to illustrate possible advanced signal processing methods that could be employed in such a framework, we perform a case study of the effect of **phase** **noise** in sensory systems that include internal oscillators. Such systems are encountered in a variety of applications, ranging from accelerometers and gy- roscopes [9], to SAW chemical sensors [10] and biochemical sensors [11]. A novel parametric analysis for the estimation of drift and **phase** **noise** variance in such sensors is presented that lifts the theoretical discrepancies of earlier analyses.

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Note that the reference **noise** is low-pass filtered and ampli- fied by the divider ratio. Although the **phase** **noise** of the reference is typically lower than that of the VCO by many orders of magnitude, it may become comparable to the VCO **noise**, if a large division factor is employed. For instance, the value of N=1024 in Winkler et al. (2005) corresponds to an increase of the **phase** **noise** by about 60 dB as it appears at the PLL output. Figure 6 shows the reference **noise** for typi- cal crystal oscillators (long dashed). Using the **noise** transfer function (dashed) we obtain the corresponding **phase** **noise** contribution (dot-dashed). Comparison with the total **phase** **noise** (solid) shows that at offsets below 1 kHz the reference **noise** dominates.

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